Foreign Object Detection Circuit Using Mutual Impedance Sensing

ABSTRACT

The present disclosure describes techniques for detecting foreign objects. In some aspects, an apparatus for detecting objects is provided. The apparatus includes a plurality of sense circuits, each of the plurality of sense circuits including a primary sense coil having a first terminal and a second terminal, a secondary sense coil having a first terminal and a second terminal, and a capacitor having a first terminal and a second terminal. The first terminal of the capacitor is electrically connected to the second terminals of each of the primary sense coil and the secondary sense coil. The apparatus further includes a driver circuit electrically connected to the first terminal of the primary sense coil of each of the plurality of sense circuits. The apparatus further includes a measurement circuit electrically connected to the first terminal of the secondary sense coil of each of the plurality of sense circuits.

RELATED APPLICATIONS

The present Application for Patent claims priority to ProvisionalApplication No. 62/783,488 entitled “FOREIGN OBJECT DETECTION CIRCUITUSING MUTUAL IMPEDANCE SENSING” filed Dec. 21, 2018 and assigned to theassignee hereof and hereby expressly incorporated by reference herein inits entirety.

FIELD

The present disclosure relates generally to object detection, forexample, in an application for inductive power transfer. In particular,the present disclosure is directed to a circuit for measuring changes ofa mutual impedance between sense coils of a plurality of sense circuits.

BACKGROUND

Object detection may be valuable for a variety of applications, and inparticular for applications where it may be useful to detect objectswithin a predetermined region. For example, in certain inductive powertransfer applications (or other types of wireless power transferapplications) it may be useful to be able to rapidly detect foreignobjects that may be present in an inductive power region and that couldbe susceptible to induction heating due to the high magnetic fieldstrength in that region. In an inductive wireless electric vehiclecharging (WEVC) system, magnetic flux densities above a transmit coil(e.g., a primary coil) can be at relatively high levels to allow forsufficient power transfer (e.g., for a WEVC system power may betransferred on the order of kilowatts—e.g., 3.3 kW, 11 kW, and thelike). Metallic objects or other objects present in the magnetic fieldcan experience undesirable induction heating. For this reason, foreignobject detection (FOD) may be implemented to detect metal objects orother objects that are affected by a magnetic field generated by thecoils of the wireless power transfer system. Solutions for improvingsensitivity, cost effectiveness, accuracy, and reliability of an objectdetection system for various applications and such as for WEVCapplications are desired.

SUMMARY

In one aspect of the disclosure, an apparatus for detecting objects isprovided. The apparatus includes a plurality of sense circuits. Each ofthe plurality of sense circuits includes a primary sense coil having afirst terminal and a second terminal, a secondary sense coil having afirst terminal and a second terminal, and a capacitor having a firstterminal and a second terminal. The first terminal of the capacitor iselectrically connected to the second terminals of each of the primarysense coil and the secondary sense coil. The apparatus further includesa driver circuit electrically connected to the first terminal of theprimary sense coil of each of the plurality of sense circuits. Theapparatus further includes a measurement circuit electrically connectedto the first terminal of the secondary sense coil of each of theplurality of sense circuits.

In another aspect of the disclosure, a method for measuring changes inelectrical characteristics for detecting objects is provided. The methodincludes applying, from a driver circuit, an input signal at anoperating frequency to a first terminal of a primary sense coil. Asecond terminal of the primary sense coil is electrically connected to afirst terminal of a capacitor. The method further includes measuring, ata measurement circuit, an electrical characteristic at an outputelectrically connected to a first terminal of a secondary sense coil. Asecond terminal of the secondary sense coil is electrically connected tothe first terminal of the capacitor. The method further includesdetecting whether an object is proximate to the primary sense coil orthe secondary sense coil based on the electrical characteristic.

In yet another aspect of the disclosure, an apparatus for detectingobjects is provided. The apparatus includes a plurality of sensecircuits. Each of the plurality of sense circuits includes a primarysense coil, a secondary sense coil, and means for compensating mutualreactance between the primary sense coil and the secondary sense coilsubstantially at an operating frequency of a sense signal. The apparatusfurther includes means for applying an input signal to each of theplurality of sense circuits. The apparatus further includes means formeasuring an output signal at an output of the secondary sense coil. Inan aspect, the apparatus may further include means for detecting whetheran object is proximate to the primary or the secondary sense coil basedon the output signal. In certain aspects, the means for detectingfurther includes means for determining a magnitude of a change in mutualimpedance between the primary sense coil and the secondary sense coilbased on the output signal.

In yet another aspect of the disclosure, an apparatus for detectingobjects is provided. The apparatus includes a plurality of sensecircuits. Each of the plurality of sense circuits includes a primarysense coil, a secondary sense coil, and a capacitor configured tocompensate mutual reactance between the primary sense coil and thesecondary sense coil substantially at an operating frequency of a sensesignal. The apparatus further includes a driver circuit electricallyconnected to the primary sense coil of each of the plurality of sensecircuits. The apparatus further includes a measurement circuitelectrically connected to the secondary sense coil of each of theplurality of sense circuits. The apparatus further includes a detectioncircuit electrically connected to the measurement circuit and configuredto determine whether an object is present based on determining amagnitude of a change in mutual impedance between the primary sense coiland the secondary sense coil based on the output of the measurementcircuit.

In yet another aspect of the disclosure, an apparatus for detectingobjects is provided. The apparatus includes a plurality of sensecircuits. Each of the plurality of sense circuits includes at least onesense coil electrically coupled to a capacitor. The apparatus furtherincludes a driver circuit including a first plurality of switches. Eachof the first plurality of switches are respectively connected to each ofthe plurality of sense circuits. The apparatus further includes ameasurement circuit comprising a second plurality of switches. Each ofthe second plurality of switches are respectively connected to each ofthe plurality of sense circuits (e.g., at an output). In some aspects,the apparatus may further include a detection circuit electricallyconnected to the measurement circuit and configured to determine whetheran object is proximate the at least one sense coil of at least one ofthe plurality of sense circuits based on an output of the measurementcircuit. In certain aspects, the driver circuit may include a currentsource circuit including an amplifier circuit having an amplifier outputand a multiplexer electrically connected between the amplifier outputand the plurality of sense circuits and configured to selectivelyconnect each of the plurality of sense circuits to the amplifier output.In certain aspects, the measurement circuit may be configured to measurea voltage at the first terminal of the secondary sense coil. In certainaspects, the driver circuit may be configured to operate as a currentsource and is configured to maintain an output wherein changes inelectrical impedance of each of the plurality of sense circuits have asubstantially negligible impact on an electrical current signal providedby the driver circuit. In certain aspects, the at least one sense coilof each of the plurality of sense circuits may be positioned to bedistributed over a predetermined area at least partially defined by awireless power transmit coil configured to inductively transfer power.In certain aspects, a frequency of a magnetic field generated by thewireless power transmit coil is different than an operating frequency ofa sense signal output by the driver circuit. In certain aspects, themeasurement circuit may include an output multiplexer formed at least inpart by the second plurality of switches and configured to selectivelyconnect each of the plurality of sense circuits to a measurementamplifier circuit. In certain aspects, each of the plurality of sensecircuits may further include a shunt inductor. In certain aspects, thecapacitor and the shunt inductor may be configured as a high pass filterconfigured to attenuate influence of a wireless power transfer field. Incertain aspects, the driver circuit may include a plurality of resistorswhere each of the plurality of resistors is respectively connected toeach of the plurality of sense circuits.

In yet another aspect of the disclosure, an apparatus for detectingobjects is provided. The apparatus includes a plurality of sensecircuits. Each of the plurality of sense circuits includes a firstprimary sense coil, a second primary sense coil, and a secondary sensecoil. The apparatus further includes a driver circuit electricallyconnected to the first primary sense coil and to the second primarysense coil of each of the plurality of sense circuits. The apparatusfurther includes a measurement circuit electrically connected to thesecondary sense coil of each of the plurality of sense circuits. In anaspect, the apparatus may include a detection circuit configured todetermine whether an object is proximate at least one of the sense coilsbased on an output of the measurement circuit. In an aspect, the drivercircuit may include a first current source circuit configured to apply afirst signal to the first primary sense coil. The driver circuit mayfurther include a second current source circuit configured to apply asecond signal to the second primary sense coil. In an aspect, themeasurement circuit may be configured to measure an open-circuit voltageat an output of the secondary sense coil. In an aspect, the firstcurrent source circuit and the second current source circuit areconfigured to control amplitudes and phases of the first signal and thesecond signal such that magnetic flux components generated by the firstprimary sense coil and the second primary sense coil are substantiallycancel (e.g., and voltage at an output of the secondary sense coil issubstantially zero or fixed in the absence of an object). In an aspect,the driver circuit may include a first input multiplexer including afirst plurality of switches electrically connected to each first primarysense coil of the plurality of sense circuits. In an aspect, the drivercircuit may include a second input multiplexer including a secondplurality of switches electrically connected to each second primarysense coil of the plurality of sense circuits. In certain aspects, thefirst primary sense coil, the second primary sense coil, and thesecondary sense coil of each of the plurality of sense circuits may bepositioned to be distributed over a predetermined area at leastpartially defined by a wireless power transmit coil configured toinductively transfer power. In certain aspects, a frequency of amagnetic field generated by the wireless power transmit coil isdifferent than an operating frequency of sense signals output by thedriver circuit.

BRIEF DESCRIPTION OF THE DRAWINGS

In the figures, the third and fourth digit of a reference numberidentify the figure in which the reference number first appears. The useof the same reference numbers in different instances in the descriptionor the figures indicates like elements.

FIG. 1 illustrates an example implementation of an object detectioncircuit.

FIG. 2A is a perspective view illustrating an example implementation ofa wireless power transfer system including the object detection circuitof FIG. 1.

FIG. 2B is a vertical cut view illustrating a portion of the wirelesspower transfer system of FIG. 2A including a portion of the objectdetection circuit of FIG. 1.

FIG. 3A is a schematic view illustrating an example of a sense coil, anobject and an associated technique based on an impedance sensingapproach that may be used in conjunction with the object detectioncircuit of FIG. 1.

FIG. 3B shows an equivalent circuit of the circuit of FIG. 3A includinga circuit model of the object.

FIG. 3C shows another equivalent circuit of the circuit of FIG. 3Aincluding the influence of the object model abstracted away.

FIG. 3D illustrates a complex impedance plane, different types ofobjects, and corresponding areas where changes of impedance may occur.

FIG. 4A is a schematic view illustrating an example of a sense coil, anobject, and an associated technique based on a capacitively compensatedimpedance sensing approach that may be used in conjunction with theobject detection circuit of FIG. 1.

FIG. 4B shows an equivalent circuit of the circuit of FIG. 4A includingthe influence of the object model abstracted away.

FIG. 5A is a schematic view illustrating an example of a double sensecoil arrangement, an object, and an associated technique based on amutual impedance sensing approach that may be used in conjunction withthe object detection circuit of FIG. 1.

FIG. 5B shows an equivalent circuit of the circuit of FIG. 5A includinga circuit model of the object.

FIG. 5C shows a ‘T’-equivalent circuit of the circuit of FIG. 5Bincluding the influence of the object model abstracted away.

FIGS. 5D to 5H are schematic cut views illustrating different exampleimplementations of the double sense coil arrangement of FIG. 5A usingplanar sense coils.

FIG. 6A is a schematic view illustrating another example of a doublesense coil arrangement, an object, and an associated technique based ona flux balanced mutual impedance sensing approach that may be used inconjunction with the object detection circuit of FIG. 1.

FIG. 6B shows a simplified ‘T’ equivalent circuit of the circuit of FIG.6A including the influence of the object model abstracted away.

FIG. 6C is a schematic view illustrating a further example of a doublesense coil arrangement, an object, and an associated technique based ona f a flux balanced mutual impedance sensing approach that may be usedin conjunction with the object detection circuit of FIG. 1.

FIG. 7A is a schematic view illustrating an example of a triple sensecoil arrangement, an object, and an associated technique based on a fluxbalanced mutual impedance sensing approach that may be used inconjunction with the object detection circuit of FIG. 1.

FIG. 7B shows an equivalent circuit of the circuit of FIG. 7A includinga circuit model of the object.

FIG. 7C shows a simplified ‘T’-equivalent circuit of the equivalentcircuit of FIG. 7B including the influence of the object modelabstracted away.

FIGS. 7D to 7H are schematic cut views illustrating different exampleimplementations of the triple sense coil arrangement of FIG. 7A usingplanar sense coils.

FIG. 8A is a schematic view illustrating an example of a double sensecoil arrangement, an object, and an associated technique based oncapacitively compensated mutual impedance sensing approach that may beused in conjunction with the object detection circuit of FIG. 1.

FIG. 8B shows an equivalent circuit of the circuit of FIG. 8A includinga circuit model of the object.

FIG. 8C shows a simplified ‘T’-equivalent circuit of the equivalentcircuit of FIG. 8B including the influence of the object modelabstracted away.

FIG. 9 is a generic block diagram illustrating example implementationsand operations of the object detection circuit of FIG. 1 using aplurality of sense circuits applicable to both impedance and mutualimpedance sensing techniques.

FIG. 10 is a circuit diagram illustrating an example implementation of aportion of the object detection circuit of FIG. 1 using a plurality ofsense circuits applicable to both an impedance sensing and a mutualimpedance sensing approach.

FIG. 11 is a circuit diagram illustrating another example implementationof a portion of the object detection circuit of FIG. 1 using a pluralityof sense circuits applicable to both an impedance sensing and a mutualimpedance sensing approach.

FIG. 12A is a block diagram illustrating example implementations of aportion of the object detection circuit of FIG. 1 using a plurality ofsense circuits applicable to the capacitively compensated mutualimpedance sensing approach of FIG. 8A.

FIG. 12B is a circuit diagram illustrating an example implementation ofthe generic circuit of FIG. 12A based on the current source voltagemeasurement approach.

FIG. 13A is a circuit diagram illustrating another exampleimplementation of a portion of the object detection circuit of FIG. 1using a plurality of sense circuits applicable to the capacitivelycompensated mutual impedance sensing approach of FIG. 8A.

FIG. 13B is a detail view illustrating a FET multiplexer switch cell ofthe circuit of FIG. 13A.

FIG. 14 is a flow diagram of an example of a method for detectingobjects.

FIG. 15 is a perspective view illustrating a wireless charging systemfor charging an electric vehicle in which any of theelements/functionality described with reference to FIGS. 1-14 may beimplemented.

FIG. 16 is a schematic view illustrating an example implementation of aportion of the wireless charging system of FIG. 15.

DETAILED DESCRIPTION

The detailed description set forth below in connection with the appendeddrawings is intended as a description of exemplary implementations andis not intended to represent the only implementations in which theinvention may be practiced. The term “exemplary” used throughout thisdescription means “serving as an example, instance, or illustration,”and should not necessarily be construed as preferred or advantageousover other exemplary implementations. The detailed description includesspecific details for the purpose of providing a thorough understandingof the exemplary implementations. In some instances, some devices areshown in block diagram form. Drawing elements that are common among thefollowing figures may be identified using the same reference numerals.

As mentioned above object detection (and particularly metal objectdetection) may be valuable for a variety of applications. For detectionin a predetermined region an object detection system may include aplurality of sense elements (e.g., a planar array of sense coils)distributed across a predetermined area. The predetermined region may bedefined by the space where metal objects may be found and where themagnetic flux density exceeds certain limits (e.g., a thresholddetermined based on what levels of temperature an object 110 might beheated up). This is generally a three-dimensional space. The number N ofsense elements may be proportional or related to the minimum size ofobjects that are desirable to be detected. For a system that isconfigured to detect small objects (e.g., the size of a paper clip), thenumber of sense elements may be relatively high (e.g., in the order of400). Drive circuitry for applying sense signals to each of the senseelements, the plurality of sense circuits each including a sense elementand additional elements for conditioning, as well as correspondingmeasurement circuitry as needed for looking for changes in electricalcharacteristics that may correspond to the presence of an object may becostly or complex as the number N of sense elements increases.

Aspects of implementations described herein relate to configurations fordriving and measurement circuitry for one or more sense circuits thatallow for higher accuracy, sensitivity, and temperature stability asneeded for detecting smaller objects. Aspects described herein relate toan implementation of the object detection circuit 100 of FIG. 1 based ona mutual impedance sensing approach that includes a sense signal sourcecharacterized by a current source (e.g., a quasi-ideal current source)that allows for determining changes in mutual impedance based onmeasuring a voltage (e.g., using a quasi-ideal voltage measurementcircuit) in one or more sense circuits. Aspects of implementations basedon mutual impedance sensing described in detail below allows forimproved detection sensitivity e.g., in presence of strong temperaturevariations such as they may be experienced in an outdoor environment.

FIG. 1 illustrates an example implementation of an object detectioncircuit 100 that includes a plurality of (N) sense circuits 104illustrated in FIG. 1 by sense circuits 104 a, 104 b, some dots, andsense circuit 104 n. The plurality of sense circuits 104 is alsoreferred herein as to the plurality of sense circuits 104 a, 104 b, . .. , 104 n. As illustrated in FIG. 1, each sense circuit of the pluralityof sense circuits 104 a, 104 b, . . . , 104 n includes a correspondingsense element (e.g., a sense coil) of a plurality of (N) sense elements106 a, 106 b, . . . , 106 n, respectively, configured to sense a changein one or more electrical characteristics that change in response to thepresence of the object 110 in proximity to at least one of the pluralityof sense elements 106 a, 106 b, . . . , 106 n to be able to providedetection coverage for an entire predetermined detection region. Each ofthe plurality sense circuit 104 a, 104 b, . . . , 104 n may also includeadditional conditioning circuitry (not shown in FIG. 1) e.g., configuredto improve measurement of a change in the one or more electricalcharacteristics. Each of the plurality of sense circuits 104 a, 104 b, .. . , 104 n also defines at least one measurement port (not shown inFIG. 1) where the change in an electrical characteristic (e.g., animpedance) is measured and refers to.

Each of the plurality of sense elements 106 a, 106 b, . . . , 106 n isshown in FIG. 1 as a “circular” coil for purposes of illustration.However, in other implementations, the sense elements 106 a, 106 b, . .. , 106 n may include a sense coil having another coil topology e.g., afigure-eight-like (DD) topology (e.g., as shown in FIG. 6C). In yetother implementations, the plurality of sense elements 106 a, 106 b, . .. , 106 n may include sense coils of a mixed coil topology e.g.,“circular” and DD (e.g., as shown in FIG. 6C). In furtherimplementations, the plurality of sense elements 106 a, 106 b, . . . ,106 n may include sense coils (e.g., solenoid coils) with a ferrite core(not shown herein) that are physically smaller compared to “air” coils.In yet further implementations, the plurality of sense elements 106 a,106 b, . . . , 106 n may include other inductive devices that can beused for generating and detecting a magnetic field for detecting anobject 110. In some implementations further described herein, each ofthe plurality of sense elements 106 a, 106 b, . . . , 106 n may includea double or even a triple sense coil arrangement (e.g., as shown inFIGS. 7F to 7H) that may be used in conjunction with a transimpedance ormutual impedance sensing approach. In some implementations, theplurality of sense elements 106 a, 106 b, . . . , 106 n is arranged inan array 106, such as a two-dimensional array 106 as shown in FIG. 1.However, in other implementations, the sense elements of the pluralityof sense elements 106 a, 106 b, . . . , 106 n are arranged in otherconfigurations that do not conform to rows or columns (radial orinterleaved), are at least partially overlapping or have irregularspacing, have different size, have different shapes (circular,hexagonal, etc.), or cover irregular detection areas, or any combinationthereof. As such the term “array” as used herein denotes a plurality ofsense elements 106 a, 106 b, . . . , 106 n that are arranged over apredetermined area. Furthermore, the number N of sense elements of anarray 106 and thus the number N of sense circuits can vary widely basedon the application including the total region in which the object 110 isto be detected and the smallest size of an object the object detectioncircuit 100 is configured to detect.

Each of the plurality of sense circuits 104 a, 104 b, . . . , 104 nincluding a corresponding sense element of the plurality of senseelements 106 a, 106 b, . . . , 106 n are operably connected to ameasurement and detection circuit 108. The measurement and detectioncircuit 108 is configured to measure one or more electricalcharacteristics at each of the plurality of sense circuits 104 a, 104 b,. . . , 104 n and process the measurements to output a signal indicativeof whether presence of an object 110 is detected. In some aspect, theoutput may include the location of the object 110 based on informationabout the particular sense element of the plurality of sense elements106 a, 106 b, . . . , 106 n at which the object 110 is detected. Themeasurement and detection circuit 108 may be configured to selectively(e.g., sequentially) apply a sense signal individually to each of theplurality of sense circuits 104 a, 104 b, . . . , 104 n to allow formeasurement of changes in at least one electrical characteristic at eachof the plurality of sense circuits 104 a, 104 b, . . . , 104 n inresponse to the presence of the object 110. The measurement anddetection circuit 108 further includes a measurement circuit to outputfor each of the plurality of sense circuits 104 a, 104 b, . . . , 104 nat least one measurement value indicative of an electricalcharacteristic.

The measurement and detection circuit 108 may include signal processingcircuitry configured to process and filter the measurement output anddetermine whether an object 110 is potentially present (e.g., based on atime-differential detection approach). At least a portion of themeasurement and detection circuit 108 may be implemented by one or moremicro-controllers or processors. For example, at least a portion of themeasurement and detection circuit 108 may be implemented as anapplication-specific integrated circuit (ASIC), a field programmablegate array (FPGA) device, digital signal processor (DSP), or anotherprocessor device. The measurement and detection circuit 108 may beconfigured to receive information from each of the components of theobject detection circuit 108 and perform calculations based on thereceived information. The measurement and detection circuit 108 may beconfigured to generate control signals for each of the components thatmay adjust the operation of that component. The measurement anddetection circuit 108 may further include a memory (not shown)configured to store data, for example, such as instructions for causingthe measurement and detection circuit 108 to perform particularfunctions, such as those related to object detection. As will bedescribed further, for purposes of selectively driving each of theplurality of sense circuits 104 a, 104 b, . . . , 104 n and measuringthe output of each of the plurality of sense circuits 104 a, 104 b, . .. , 104 n further analog or other circuit components may be included inthe measurement and detection circuit 108.

In an exemplary implementation, an object 110 is detected by applying asinusoidal sense signal and by measuring a change of an impedance ateach of the plurality of sense circuits 104 a, 104 b, . . . , 104 n.While the description may refer to sinusoidal signals it should beappreciated that any alternating voltage or alternating current may beapplied and are contemplated by different implementations describedherein. For example, the object detection circuit 100 may be configuredto detect metallic objects (or other objects) that can cause changes inat least one of an impedance (e.g., a mutual impedance) as measured ineach of the plurality of sense circuits 104 a, 104 b, . . . , 104 n. Inan exemplary implementation, the measurement and detection circuit 108is configured to cause each of the plurality of sense elements (e.g.,sense coils) 106 a, 106 b, . . . , 106 n (e.g., sequentially) togenerate an alternating magnetic field at an operating frequency alsoreferred herein as to the sense frequency f_(s). If a metallic object110 is present in the alternating magnetic field as generated by a senseelement, eddy currents will be generated in the object 110 if the objectis electrically conductive (metallic). According to Lentz' law, the eddycurrents in the object 110 will generate another (secondary) magneticfield that interacts with the respective sense element (e.g., a mutualcoupling is developed). This may cause a change in an electricalcharacteristic (e.g., an impedance) of at least a portion of theplurality of sense elements 106 a, 106 b, . . . , 106 n and thus achange in an electrical characteristic as measured at the at least onemeasurement port of the corresponding sense circuits of the plurality ofsense circuits 104 a, 104 b, . . . , 104 n. Other interactions such aselectric field (capacitive) interactions or ferromagnetic interactionbetween an object 110 and a sense element (e.g., a sense coil) of theplurality of sense elements 106 a, 106 b, . . . , 106 n are alsopossible that cause a change in an electrical characteristic (e.g., animpedance) of at least a portion of the plurality of sense elements 106a, 106 b, . . . , 106 n.

In other implementations, an object 110 is detected by applying a sensesignal different from a sinusoidal signal (single frequency) and bymeasuring a change in a response to that signal. In an exemplaryimplementation, the measurement and detection circuit 108 is configuredto selectively (e.g., sequentially) excite each of the plurality ofsense elements 106 a, 106 b, . . . , 106 n with a pulse suitable formeasuring an impulse response and presence of an object is determinedbased on measuring a change in an impulse response in each of theplurality of sense circuits 104 a, 104 b, . . . , 104 n.

In another exemplary implementation, the measurement and detectioncircuit 108 is configured to selectively (e.g., sequentially) drive eachof the plurality of sense circuits 104 a, 104 b, . . . , 104 n with amultiple frequency (multi-tone) signal and presence of an object isdetermined based on measuring a change in an impedance as measured ineach of the plurality of sense circuits 104 a, 104 b, . . . , 104 n andfor each frequency component individually.

In another exemplary implementation, the measurement and detectioncircuit 108 is configured to selectively (e.g., sequentially) drive eachof the plurality of sense circuits 104 a, 104 b, . . . , 104 n withanother suitable waveform (e.g., a pseudo-noise signal) and presence ofan object 110 is determined based on measuring a change in a response tothat waveform in each of the plurality of sense circuits 104 a, 104 b, .. . , 104 n.

The descriptions of the object detection circuit 100 herein assume ameasurement and detection circuit 100 that is configured to detectchanges in an impedance at a single frequency and determine if theobject 110 is present in proximity to at least one of the plurality ofsense elements 106 a, 106 b, . . . , 106 n. However, this should notexclude implementations using a measurement and detection circuit 100that is configured to detect changes in one or more electricalcharacteristics using other sense signal waveforms to determine whetheran object 110 is present.

Example Inductive Wireless Power Application for Object Detection

The object detection circuit 100 of FIG. 1 may be used in a variety ofapplications for detecting objects, such as metallic objects, within apredetermined detection region as previously defined. In one examplementioned above, in an inductive wireless power transfer system,magnetic flux densities above a transmit coil (e.g., a primary coil) andbelow a receive coil (e.g., secondary coil) can be at relatively highlevels to allow for sufficient power transfer (e.g., for a wirelesselectric vehicle charging (WEVC) system power may be transferred on theorder of kilowatts, e.g., 3.3 kW, 11 kW, or at even higher levels).Metallic objects or other objects present in the magnetic field canexperience undesirable induction heating based on interaction with thewireless power field. For this reason, the object detection circuit 100may be integrated into a wireless power transfer system to detect metalobjects or other objects that are affected by a magnetic field generatedby the coils used for the wireless power transfer. Such detection mayallow the wireless power transfer system to respond appropriately (e.g.,reduce or stop power transmission, alert a user, and the like).

FIG. 2A illustrates an example implementation of a wireless powertransfer system 200 including the object detection circuit 100 of FIG.1, a power conversion circuit 222 and a wireless power transferstructure 224. The wireless power transfer system 200 may depict eithera wireless power transmit system that generates a magnetic field 232 fortransferring power or a wireless power receive system that can coupleand receive power via the magnetic field 232. When the wireless powertransfer system 200 is configured as a wireless power transmit system,the power conversion circuit 222 is configured to convert power from apower source (not shown) to a suitable operating frequency (e.g., 85kHz) and form for wireless power transfer via the wireless powertransfer structure 224. It may be more likely that when integrated withan object detection circuit 100, the wireless power transfer system 200may be a wireless power transmit system as power may be generallytransferred from the ground or other upward facing surface wheremetallic objects will generally come to a rest. However otherimplementations are possible, e.g., the object detection circuit 100 ora portion thereof may be also integrated into a wireless power receivesystem. When the wireless power transfer system 200 is configured as awireless power receive system, the power conversion circuit 222 isconfigured to convert power received via the wireless power transferstructure 224 into a suitable form (e.g., DC with a suitable voltage andcurrent level) for providing power to a load, such as a battery.

The wireless power transfer structure 224 (also sometimes referred to asa “pad”) is configured to wirelessly transmit or receive power. FIG. 2Aillustrates one example of a wireless power transfer structure 224 andhow the sense element array 106 of FIG. 1 may be integrated. Thewireless power transfer structure 224 includes a coil 226 also referredto as the wireless power transfer coil that is configured to generate analternating magnetic field when driven with a current by the powerconversion circuit 222. The wireless power transfer structure 224 mayfurther include ferrite 228 configured to channel and/or provide a pathfor magnetic flux (e.g., may be arranged in one or more ferrite barswhich can be a combination of ferrite tiles arranged to form the bars).The power transfer structure 224 may also include a shield 230 (alsosometimes referred to as a back plate). The shield 230 is configured toprevent the magnetic field 232 or associated electromagnetic emissionsfrom extending beyond a boundary determined by the shield 230 or atleast to attenuate the magnetic field 232 extending beyond thatboundary. As an example, the shield 230 may be formed from aluminum.

In the illustrated example, the wireless power transfer structure 224includes a double-D (DD) coil topology, which includes two electricallyconductive structures disposed proximate to one another forming the DDwireless power transfer coil 226. The wireless power transfer structure224 is configured to generate a magnetic field (indicated in FIG. 2A byline of flux 232) by running alternating current through the DD wirelesspower transfer coil 226. Generally, the current in the two centersections of the DD wireless power transfer coil 226 runs in the samedirection. In this way, a high magnetic flux is generated in the centerof the DD wireless power transfer coil 226 and is channeled through theferrite 228 and arches above the DD wireless power transfer coil 226from one opening (magnetic pole area) to the other opening (magneticpole area) of the DD wireless power transfer coil 226 as indicated inFIG. 2A by a line of flux. Other coil topologies are also applicable tothe techniques described herein, including a multi-coil topology (e.g.,Bi-polar, DD plus Circular) or just a single coil Circular or Solenoidtopology.

The wireless power transfer system 200 further includes the objectdetection circuit 100 of FIG. 1 that may include a plurality of sensecircuits 104 a, 104 b, . . . , 104 n each sense circuit including asense element (e.g., a sense coil) of the plurality of sense elements106 a, 106 b, . . . , 106 n as illustrated in FIG. 2A. The plurality ofsense elements 106 a, 106 b, . . . , 106 n may be arranged in asubstantially planar array 106 to cover a predetermined area (e.g., atleast the area covered by the wireless power transfer coil 226 or theferrite 228). If each sense element of a plurality of sense elements 106a, 106 b, . . . , 106 n constitutes a sense coil (or an arrangement ofsense coils), then the size and the number N of sense elements 106 a,106 b, . . . , 106 n may depend on the size of the wireless powertransfer coil 226 or the ferrite 228 and also on the smallest size of anobject 110 that is required to be detected by the object detectioncircuit 100. For example, if the minimum size of a metallic object 110required to be detected is the size of a coin (e.g., a 1 €cent coin),then the size of each sense element of the plurality of sense elements106 a, 106 b, . . . , 106 n may be either on the order of this minimumsize or a multiple thereof (e.g., with a 10-times larger area), assumingthat those minimum size objects are located in close proximity of aplane defined by the array 106 (e.g., on the top surface of the housing236 of the base pad as shown later in FIG. 2B). As a mere illustrativeexample, the number N of sense elements of the array 106 could be on theorder of 64 (e.g., 8×8 array) to be able to provide coverage of theentire predetermined area with the required sensitivity.

Further the system shown in FIG. 2A may include a housing (not shown inFIG. 2A but shown later in FIG. 2B as housing 236) configured to house,for example at least the wireless power transfer coil 226, the ferrite228, the sense element array 106, and potentially the shield 230. Thehousing may be made of any suitable material (e.g., hard plastic,ceramics etc.) and can be designed to provide structural support forexample to support the weight of various objects such as vehicles thatmay pass over the housing. In some implementations, the housing may bemade of a non-conductive material to avoid interfering or interactingwith the magnetic field 232. The sense element array 106, in oneexemplary implementation, is positioned between the wireless powertransfer coil 226 and the housing 236 (with other intervening layers ifdesired) so that the sense elements of the plurality of sense elements106 a, 106 b, . . . , 106 n are positioned closer to objects that mayrest on the surface of the housing 236 and where magnetic field levelscould be high during power transfer.

All or just a portion of the power conversion circuit 222 may also behoused in the housing 236. Although in some implementations the powerconversion circuit 222 may be housed separately from the housing 236that houses the wireless power transfer structure 224. In some cases,the power conversion circuit 222 is housed in the housing 236 but ispositioned on the other side of the shield 230 from the ferrite 228.

FIG. 2B illustrates a vertical cut view of a portion 250 of a wirelesspower transfer system 200 with reference to FIG. 2A and applicable to aWEVC application. This portion 250 includes the base-side (e.g.,transmit) wireless power transfer structure 224 and the vehicle-side(e.g., receive) wireless power transfer structure 260. The base-sidewireless power transfer structure 224 includes a shield (back plate) 230made of an electrically conductive material, a layer of ferrite 228 anda wireless power transfer coil 226. It also includes a housing 236configured to house the wireless power transfer coil 226, ferrite 228,and shield 230. In addition, the housing 236 is configured to house asense element array 106 as part of the object detection circuit 100 asillustrated in FIG. 2A. In some implementations, the shield 230 may forma portion of the housing 236. The power conversion circuit 222 is notshown but may be electrically connected to the wireless power transfercoil 226 or a portion or all may also be housed in the housing 236.

The vehicle-side wireless power receive structure 260 includes awireless power transfer coil 266, a layer of ferrite 268, and a shield270 made of an electrically conductive material. In someimplementations, the shield 270 may be formed from a portion of theapparatus that the ferrite 268 and the wireless power transfer coil 266are affixed to (e.g., the metallic underbody of a vehicle). In thiscase, a housing 276 configured to house the wireless power transfer coil266 and ferrite 268 is provided but that would not house the shield 270.However other implementations are possible where a shield 270 (e.g., aback plate) is included in the housing 276. A power conversion circuit222 is not shown but may be electrically connected to the receive coil268 or a portion or all may also be housed in the housing 276.

The base-side wireless power transfer structure 224 may be configured togenerate a magnetic field 232. The vehicle-side wireless power structure260 may be configured to inductively receive power via the magneticfield 232. Magnetic flux 232 may be at a particular level (flux density)at the surface of the housing 236. In some cases, flux density may besomewhat higher at or in proximity to the top surface of the housing 236relative to surrounding areas as the surface of the housing 236 iscloser to the wireless power transfer structure 226. Furthermore, as thewireless power transfer structure 224 may be positioned on a ground orother top facing surface, an object 110 may come to rest at the topsurface of the housing 236 as illustrated in FIG. 2B. The object 110 maythereby be potentially exposed to high levels of magnetic flux densityif power is being transferred. The object detection circuit 100 isconfigured to detect the object 110 using the sense element array 106.

Inductive Object Detection Techniques

FIGS. 3 to 8 illustrate examples of different techniques based onmeasuring at least one electrical characteristic that may be used inconjunction with the object detection circuit 100 of FIG. 1. Theseexamples are to illustrate the principle of the sensing and measurementtechnique and do not show all the details of an object detection circuit100. Particularly, they do not show the further signal processing andevaluation circuit as it may be required e.g., for detecting an objectbased on a change in a measured electrical characteristic. Forillustrative purposes and simplicity, these techniques are illustratedby means of a single sense element though applicable to a plurality ofsense elements (e.g., sense elements 106 a, 106 b, . . . , 106 n) asdescribed below with reference to FIGS. 10 to 14.

FIG. 3A shows a circuit 300 to illustrate a technique for inductivelysensing a presence of an object 110 based on a change in a compleximpedance Z₁ as measured at the terminals of sense coil 302 (e.g., aplanar a multi-turn coil) with inductance L₁ that may represent e.g.,sense element 106 a. This technique is also referred herein as to theimpedance sensing approach. A change ΔZ₁ of the impedance Z₁ relative tothe impedance Z_(1,0) in the absence of the object 110 may indicate apresence of the object 110. A change of impedance may be also producedwhen the sense coil 302 is integrated into the wireless power transferstructure 224 as shown in FIG. 2A due to e.g., electrically conductivematerials, ferromagnetic materials, but also due to dielectric materials(e.g., plastic housing 236) that may be located in the proximity of thesense coil 302. Presence of such materials is indicated in FIG. 3A bymaterials 310 (shaded area). Materials 310 may also include a dielectricsubstrate that carries the sense coil 302 or a dielectric medium thesense coil is embedded in (e.g., in case of a printed circuit board(PCB) design). The effects of materials 310 may generally produce achange of the sense coil's 302 inductance, resistance, but also of itsself-capacitance relative to its inductance, resistance, andself-capacitance as measured in free space. The effects of materials 310are considered already included in the impedance Z_(1,0).

The circuit 300 of FIG. 3A illustrates a technique for measuring animpedance Z₁ where a sinusoidal current I₁ with a defined frequency,amplitude, and phase provided by a current source 306 is applied to thesense coil 302 and where the open-circuit voltage V₁ across theterminals of the sense coil 302 is measured. The open-circuit voltage V₁is measured using a sensitive high impedance voltage measurement circuit304 so that there is virtually zero current at the terminals of thevoltage measurement circuit 304 (I₂≅=0).

This impedance measurement technique is also referred herein as to thecurrent source voltage measurement approach. In some implementations,the current source 306 and the voltage measurement circuit 304 may bepart of the measurement and detection circuit 108 with reference toFIG. 1. The voltage measurement circuit 304 may be frequency selective(narrowband) tuned to the sense frequency f_(s) (frequency of thecurrent source) which may be e.g., in the MHz range. The impedanceZ₁=Z_(1,0)+ΔZ₁ of the sense coil 302 may be determined by dividing themeasured voltage V_(1,0)+ΔV₁ by the defined (known) current I₁, whichmay be expressed more formally as

$\begin{matrix}{Z_{1} = {{Z_{1,0} + {\Delta Z_{1}}} = {\frac{V_{1,0} + {\Delta V_{1}}}{I_{1}}.}}} & (1)\end{matrix}$

Using an ideal current source 306, a change ΔZ₁ in the impedance Z₁e.g., due to presence of the object 110 manifests in a change ΔV₁ in thevoltage V₁ while the current I₁ remains unaffected. Therefore, measuringa change ΔV₁ in the voltage V₁ may be equivalent to measuring a changeΔZ₁ in the impedance Z₁. In other words, the voltage V₁ may beindicative of the impedance Z₁.

Though not shown herein, other impedance measurement techniques are alsocontemplated e.g., by applying a sinusoidal voltage source with adefined voltage V₁ (amplitude and phase) and measuring the currentI_(1,0)+ΔI₁ at the sense coil 302 using a sensitive (e.g., frequencyselective) low impedance current measurement circuit. This impedancemeasurement technique is also referred herein as to the voltage sourcecurrent measurement approach.

The current source 306 used in an implementation of the current sourcevoltage measurement approach may be characterized by a quasi-idealcurrent source. A quasi-ideal sinusoidal current source may be definedas a sinusoidal signal source with a sufficiently large (but finite)source impedance so that the magnitude of the fractional change in itsoutput current |ΔI₁/I_(1,0)| is at least a factor of 10 smaller than themagnitude of the fractional change in the voltage |ΔV₁/V_(1,0)|, whereΔI₁ and ΔV₁ denote the change in the complex amplitude (magnitude andphase) in its output current I_(1,0) and in the voltage V_(1,0)respectively, due to presence of the object 110, where I_(1,0) andV_(1,0) refer to the complex amplitude of its output current and thevoltage, respectively, in absence of the object 110. More formally, thisdefinition may be expressed by the following inequality

$\begin{matrix}{| \frac{\Delta V_{1}}{V_{1,0}} \middle| \middle| \frac{I_{1,0}}{\Delta \; I_{1}} \middle| {\geq {10}} ,} & (2)\end{matrix}$

This definition (Equation (3)) may apply to the current source (e.g.,current source 306) as it may be used in implementations based on any ofthe impedance or mutual impedance sensing techniques described herein.In any case, the voltages V_(1,0) and ΔV₁ in Equation (3) refer to therespective voltages across the output of the current source (e.g.,current source 306) and the currents I_(1,0) and ΔI₁ to the respectivecurrents at the output of the current source.

Likewise, the voltage source in an implementation of the voltage sourcecurrent measurement approach may be characterized by a quasi-idealvoltage source. A quasi-ideal sinusoidal voltage source may be definedas a sinusoidal signal source with a sufficiently small (but non-zero)source impedance so that the magnitude of the fractional change of itsoutput voltage |ΔV₁/V_(1,0) is at least a factor of 10 smaller than themagnitude of the fractional change of its output current |ΔI₁/I_(1,0)|,where ΔV₁ and ΔI₁ denote the change in complex amplitude (magnitude andphase) of its output voltage and current, respectively, due to presenceof an object 110 and V_(1,0) and I_(1,0) the complex amplitude of itsoutput voltage and current, respectively, in absence of the object 110.

Above definitions may be generalized to non-sinusoidal signal (arbitrarywaveform) sources, where the notions of complex impedance and complexamplitude may not directly apply. This may be accomplished byapproximating the signal by a complex Fourier series and applying abovedefinitions to the individual frequency components of the complexFourier series.

The voltage measurement circuit 304 used in an implementation may becharacterized by a quasi-ideal voltage measurement circuit 304 whoseinput impedance magnitude is sufficiently large such that the magnitudeof the measurement error E (complex value) as produced by thequasi-ideal voltage measurement circuit 304 is less than 10% of themagnitude of the voltage V_(1,∞) as measured with an ideal voltagemeasurement circuit 304 (infinite input impedance). More formally, thisdefinition may be expressed as

$\begin{matrix}{{\frac{|ɛ|}{| V_{1,\infty} |} = {\frac{| {V_{1} - V_{1,\infty}} |}{| V_{1,\infty} |} < 0.1}},} & (3)\end{matrix}$

where V₁ refers to the voltage as measured with the quasi-ideal voltagemeasurement circuit 304 (finite input impedance). This definition(Equation (5)) may apply to the voltage measurement circuit (e.g.,voltage measurement circuit 304) as it may be used in implementationsbased on any of the impedance and mutual impedance sensing techniquesdescribed herein. In any case, the voltages V₁ and V_(1,∞) in Equation(5) refer to the respective voltages across the input of the voltagemeasurement circuit (e.g., voltage measurement circuit 304).

An equivalent circuit of the circuit 300 of FIG. 3A including a circuitmodel of the object 110 is shown in FIG. 3B. In this equivalent circuit,the sense coil 302 is represented by an equivalent inductance L₁(ϑ) andan equivalent loss resistance R₁(ϑ) both shown as a function oftemperature ϑ. As previously stated, inductance L₁(ϑ) may include achange of an inductance due to the presence of materials 310. Dependingon the electromagnetic properties of materials 310, this may be adecrease or increase of the inductance L₁(ϑ) as measured in free space.

The loss resistance R₁(ϑ) may include two resistance components. A firstresistance component may be attributed to the resistance of the sensecoil's conductive structure (e.g., copper wires or PCB traces) asexperienced at the sense frequency f_(s) (subject of skin and proximityeffects). A second resistance component may be due to loss effects inmaterials 310 (e.g., eddy current and/or hysteresis losses). It may beappreciated that both resistance components may be subject of thermaleffects, e.g., they may increase if the temperature inside the housing236 of the wireless power transfer structure 224 rises. As an example,the DC resistance of a sense coil's 302 copper winding may vary by 40%for a temperature variation over the range from −20° C. to +80° C. as itmight be specified for an installation in an outdoor environment andassuming a temperature coefficient of 0.004Ω/K for copper and a linearrelationship. This resistance variation may be smaller (in the order of20%) at sense frequency f_(s) (e.g., in the MHz range), taking skin andproximity effects into account, since current distribution inside thesense coil's 302 conductive structure (e.g., wire) also changes with theelectrical conductivity and thus with temperature. A similar variationmay be expected in the second resistance component if eddy currentlosses in the materials 310 are the predominant loss effect. Forcomparison, the percental change |ΔZ₁|/R₁(ϑ) produced by a small object110 (e.g., a paper clip) may be in the order of only 0.1% assuming acost optimized implementation of the object detection circuit 100 ofFIG. 1 providing a plurality of sense coils 106 a, 106 b, and 106 n eachwith a form factor in the order of 60×80 mm. This example demonstratesthat significant changes in impedance Z over time may occur if theobject detection circuit 100 is operated in an outdoor environment.Given such magnitudes of thermal effects, object detection may require adifferential detection approach (e.g., a time differential detectionscheme) rather than detection on an absolute basis (absolute detection)e.g., by comparing the measured impedance Z₁ against a reference valueZ_(1,ref) that has been determined in a process of calibration e.g., attime of installation or commissioning of the wireless power transferstructure 224 as further discussed below.

While significant thermal drift may be expected in the equivalent lossresistance R₁(ϑ) (real part of impedance Z₁), there may be also thermaleffects in the equivalent inductance L₁(ϑ) (imaginary part of impedanceZ₁ divided by 2 πf_(s)) of the sense coil 302 as measured at sensefrequency f_(s). Variations in the equivalent inductance L₁(ϑ) may bedue to a change of the sense coil's inner inductance that may alsochange with temperature ϑ since electrical conductivity (e.g., theconductivity of copper) and consequently the current distribution (skinand proximity effects) inside the sense coil's 302 wire may change withtemperature. Thermal variations of the equivalent inductance L₁(ϑ) mayalso emanate from thermal expansion e.g., of the PCB carrying sense coilarray 106 and also due to micro-mechanical movements of the sense coilarray 106 relative to materials 310 due to thermal expansion. Further,they may be produced by a temperature dependent permeability ifmaterials 310 include a ferromagnetic material (e.g., ferrite) and alsoby a temperature dependent permittivity if materials 310 include adielectric material (e.g., FR4 PCB substrate) affecting the sense coil's302 self-capacitance. Such thermal capacitive effects may becomesignificant in certain implementations of the object detection circuit100 of FIG. 1 operating at frequencies f_(s) in the MHz range.

In FIG. 3B, the object 110 is modelled by an equivalent inductance L₃and an equivalent loss resistance R₃ justified by the fact that anobject 110 in general can store and dissipate electrical energy. AnLR-model may apply to a metallic object 110 that appears electricallyconductive but non-ferromagnetic at sense frequency f_(s). It may notapply to an object 110 that appears ferromagnetic or dielectric at sensefrequency f_(s) as further discussed below. An object 110 not exhibitinga noticeable ferromagnetic effect (e.g., a magnetic relativepermeability μ_(r)>1) at sense frequency f_(s) may be referred to as anon-ferromagnetic object. Conversely, an electrically conductive object110 that appears ferromagnetic at sense frequency f_(s) is referred toas a ferromagnetic conductive object 110. Modelling of a ferromagneticconductive object 110 may be more complex than shown in FIG. 3B by anequivalent LR circuit model. Magnetic coupling between object 110 andsense coil 302 (between inductance L₃ and inductance L₁) is modelled bycoupling factor k₁₃. Practical experience with many different types ofnon-ferromagnetic objects 110 shows that variations of position andorientation of the object 110 relative to sense coil 302 mainly affectthe coupling factor k₁₃ but generally have a minor impact on theparameters of its LR model. Therefore, as a first approximation, anon-ferromagnetic object 110 may be modelled with fixed parameters forL₃ and R₃ regardless of its position and orientation. Both resistance R₃and inductance L₃ may also be functions of the object's 110 temperaturethough not indicated in FIG. 3B. Thermal effects in the object 110 maybe less relevant, except in certain implementations of the objectdetection circuit 100 of FIG. 1 capitalizing on a temperature dependencein R₃ and L₃ e.g., by detecting objects based on a correlation with thelevel of magnetic field of the wireless power transfer (e.g., atf_(wpt)=85 kHz) that inductively heats the object 110 as describedfurther below in more detail.

Defining the angular sense frequency

ω_(s)=2πf _(s),  (4)

the impedance of the sense coil 302 at sense frequency f_(s) in absenceof the object 110

Z _(1,0) =R ₁ +jω _(s) L ₁,  (5)

the impedance of the object 110

Z ₃ =R ₃ +jω _(s) L ₃,  (6)

and the mutual inductance between object 110 and sense coil 302

M ₁₃ =k ₁₃√{square root over (L ₁ L ₃)},  (7)

the impedance Z₃ as measured at the terminals of the sense coil 302 inpresence of object 110 may be expressed as

Z ₁ =Z _(1,0) +ΔZ ₁ =Z _(1,0)+α₁₃ ² Z ₃*,  (8)

with

$\begin{matrix}{{\alpha_{13} = \frac{\omega_{s}M_{13}}{| Z_{3} |}},} & (9)\end{matrix}$

where α₁₃ denotes a transformation factor and Z₃* the conjugate complexof Z₃.

Equations (8) and (9) show that the equivalent circuit model of theobject 110 in FIG. 3B may be abstracted away as an impedance (impedancechange)

ΔZ ₁=α₁₃ ² Z ₃*,  (10)

in series to the sense coil's equivalent circuit (L₁(ϑ), R₁(ϑ)) asillustrated in FIG. 3C. Assuming a scalar (non-complex) coupling factork₁₃, the transformation factor α₁₃ is also a scalar and the change inimpedance ΔZ₁ reflects the conjugate complex of the impedance of theobject 110 with respect to the Q-factor or the angle (argument) of theimpedance Z₃. A Q-factor may be attributed to the object 110 since theobject 110 in general can store and dissipate energy. Eddy currents areinduced into a metallic object 110 when subjected to the magnetic sensefield as generated by sense coil 302. Energy is stored in the secondarymagnetic field produced by the induced eddy currents in the object 110(inductance L₃) and dissipated in its resistance R₃. The Q-factor of theobject 110 may be defined as

$\begin{matrix}{Q_{3} = {\frac{\omega_{s}L_{3}}{R_{3}}.}} & (11)\end{matrix}$

Likewise, a Q-factor may be attributed the impedance change ΔZ₁(reflected impedance of the object 110) defined as

$\begin{matrix}{Q_{\Delta \; Z_{1}} = \frac{{Im}\{ {\Delta \; Z_{1}} \}}{{Re}\{ {\Delta Z_{1}} \}}} & (12)\end{matrix}$

where Re{·} and Im{·} denote the real and imaginary part. The relationbetween Q_(Δz) ₁ and Q₃ may be expressed as

Q _(Δz) ₁ =−Q ₃.  (13)

The Q-factor of ΔZ₁ equals the sign inverted Q-factor of the object 110.Alternatively, using the angles arg{·} of the impedances, this relationmay be expressed as

arg{ΔZ ₁}=−arg{Z ₃}.  (14)

Equation (10) indicates that presence of a non-ferromagnetic metallicobject 110 with finite electrical conductivity, which can be modelledwith resistance R₃>0 and inductance L₃>0, produces a positive Re{ΔZ₁}and a negative Im{ΔZ₁}. Otherwise stated, it produces an increase ofequivalent resistance and a decrease (destruction) of equivalentinductance in the circuit 300.

FIG. 3D illustrates a complex plane 330 or more precisely a complex halfplane comprising quadrant 1 and 4 where the impedance changes ΔZ₁(responses) of different types (categories) of objects 110 may occur.More particular, FIG. 3D shows shaded areas (angle ranges) correspondingto different categories of objects 110 where ΔZ₁ may be measured at asense frequency f_(s) (e.g., in the MHz range) if the object 110 isplaced in proximity of the sense coil 302. To emphasize thecharacteristics of the different categories of objects 110, the angleranges indicated in FIG. 3D may be not drawn to scale and should beconsidered qualitative rather than quantitative. The actual angle rangesmay also depend on the particular sense frequency f_(s). Since theQ-factor of some categories of object 110 generally increases withfrequency, some areas will move closer to the imaginary axis when thesense frequency f_(s) is increased.

A non-ferromagnetic object 110 with a well conducting surface (e.g., acopper coated coin with a coating equal or thicker than the eddy currentpenetration (skin) depth δ) may produce a ΔZ₁ in the angle range 331close to the negative imaginary axis in the 4^(th) quadrant of thecomplex plane 330 indicating an object 110 with a relatively highQ-factor Q₃.

The angle range 332 that is also in the 4^(th) quadrant may becharacteristic for a piece of thin foil or a metallized (aluminumcoated) paper. Such non-ferromagnetic objects 110 may exhibit a lowerQ-factor Q₃ than e.g., a copper coated coin. This may be particularlytrue, if the electrical conductivity a of the metal (e.g., aluminum)coating is lower than that of copper and if the thickness of the coatingis smaller than the theoretical skin depth

$\begin{matrix}{\delta = \sqrt{\frac{1}{\mu_{0}\mu_{r}\sigma f_{s}}}} & (15)\end{matrix}$

where μ₀ denotes the magnetic permeability constant and μ_(r) therelative permeability that is one.The angle range 333 may be typical for the response ΔZ₁ of someferromagnetic steel objects (e.g., nuts). These objects 110 areelectrically conductive but also exhibit a relative permeabilityμ_(r)>1. The effect of ferromagnetism in these objects 110 may bethreefold. First, it may increase the reactance Im{Z₁} of the sense coil302. Second, it may increase the loss resistance Re{Z₁} compared to anequal but non-ferromagnetic object with the same conductivity σ sinceskin depth δ reduces thus resistance R₃ increases with increasingpermeability μ_(r) as evident from equation (15). Third, since theobject 110 is conductive, it may also destroy reactance of the sensecoil 302 at the same time. Therefore, the net response produced by someferromagnetic steel objects 110 (e.g., nuts) may be found close to thereal axis in the angle range 333 (e.g., Im{ΔZ₁}≅0) in the 4^(th) or inthe 1^(st) quadrant of the complex plane 330. With an implementationthat detects objects 110 e.g., solely based on a change of reactanceIm{ΔZ₁}, such objects 110 may appear as stealth objects 110.

Ferromagnetic steel objects 110 with a cylindrical shape of diameter dmuch smaller than its length l (e.g., nails, pins, pieces of steel wire)may produce a response ΔZ₁ in the angle range 334 around 45° in the1^(st) quadrant somewhat depending on its orientation relative to themagnetic sense field. For this category of objects 110, theferromagnetic effect increasing a reactance Im{Z₁} may be much largerthan the reactance destroying effect of its conductivity. These objects110 may produce an impedance change ΔZ₁ with a positive Q-factor Q_(Δz)₁ close to unity.

Ferromagnetic non-conductive objects 110 with low resistive losses(e.g., a piece of ferrite) may produce a response ΔZ₁ in the angle range335 close to the positive imaginary axis corresponding to a highpositive Q-factor Q_(Δz) ₁ .

A similar response ΔZ₁ in the angle range 335 may be also produced bydielectric non-conductive objects 110 with low resistive losses (e.g., ahuman hand, a plastic bottle filled with water). Dielectric objects 110may interact with the sense coil 302 via the electric field generated bythe sense coil's 302 self-capacitance that may be modelled by acapacitance C_(1,self) in parallel to the equivalent circuit of thesense coil 302 (not shown in FIG. 3B but considered merged into theequivalent inductance L₁). A dielectric non-conductive object 110 withlow resistive losses in proximity of the sense coil 302 may generallyincrease self-capacitance C_(1,self) resulting in an increase ofreactance (Im{ΔZ₁}>0) as evident from the following equation:

L ₁ ≅L _(1,ex)+ω_(s) ² L _(1,ex) ² C _(1,self),  (16)

where L_(1,ex) denotes the sense coil's 302 inductance excludingself-capacitance.

In some aspects of an object detection circuit 100, the sense coil 302may be used for capacitive sensing of living objects e.g., a human hand,a cat, or any other animal that are predominantly dielectric and thatmay be located in proximity of sense coil 302. Such use case may requirethe object detection circuit 100 to discriminate dielectric objects frommetallic objects e.g., if rules and procedures for living objectdetection would differ from those applied to metal object detection.

Objects 110 producing a response ΔZ₁ in angle ranges 332, 333, and 334may be subject of significant induction heating if exposed to the strongmagnetic field of the wireless power transfer (e.g., at f_(wpt)=85 kHz)due to their losses (low Q-factor). This may be particularly true, forthe cylindrical ferromagnetic objects 110 that may produce an impedancechange in the angle range 334. Ferromagnetism of steel objects 110 maysaturate e.g., at a r.m.s. magnetic flux density level of 1 mT. For fluxdensities above that level, excessive hysteresis losses and consequentheating effects may occur. This object category may be characterized bythe highest loss power density (e.g., Watt per unit surface area) andthus highest temperature. Therefore, in some aspects of an objectdetection circuit 100, it may be desirable to selectively increase asensitivity to objects 110 of this category.

To discriminate between certain categories objects 110 and/or increase asensitivity for certain categories of objects 110, in an aspect theobject detection circuit 100 may be configured to measure an impedancechange ΔZ₁ with sufficient accuracy at least with respect to its anglearg{ΔZ₁}. Even higher accuracy (e.g., angle fidelity) may be required ifan object detection circuit 100 employs a time-differential detectionscheme as further described in connection with FIG. 9. A fast (e.g.,abrupt) change ΔZ₁ in a sequence (time-series) of consecutively measuredimpedances Z₁ due to the object 110 brought to proximity of sense coil302 may cause a time-differential detector to temporarily produce anoutput indicative for ΔZ₁. In contrast, a fast (e.g., abrupt) change ΔZ₁due to the object 110 removed from the proximity of sense coil 302 maycause the time-differential detector to temporarily produce an outputindicative for −ΔZ₁ (opposite sign). Therefore, outputs of atime-differential detector may fall in all four quadrants of the complexplane 330, depending on the characteristics (e.g., impedance) of theobject 110 and whether it is brought to or removed from the proximity ofthe sense coil 302. In some aspects of an object detection circuit 100,it may be desirable to discriminate between an object 110 entering thepredetermined space and an object 110 leaving this space. Therefore,some implementations of an object detection circuit 100 of FIG. 1 may beconfigured to provide accurate calibration of the impedance measurementat least with respect to its angle arg{ΔZ₁}. This may be particularlytrue for implementations of the object detection circuit 100 of FIG. 1relying on an impedance measurement circuit (e.g., including aquasi-ideal current source 306 and a quasi-ideal voltage measurementcircuit 304) that is subjected to measurement errors.

In some other aspects of the object detection circuit 100 of FIG. 1, theobject detection sensitivity may be defined as the impedance change ΔZ₁as produced in presence of an object 110 normalized to |Z_(1,0)| that isthe magnitude of the impedance as measured in absence of the object 110.This normalized impedance change (sensitivity) is herein also referredto as the fractional change ΔZ₁/|Z_(1,0)|. For a non-ferromagneticobject 110, the fractional change may be expressed as

$\begin{matrix}{\frac{\Delta Z_{1}}{| Z_{1,0} |} = {\frac{\alpha_{13}^{2}Z_{3}^{*}}{| Z_{1,0} |}.}} & (17)\end{matrix}$

For a sense coil 302 with high enough (native) Q-factor

$\begin{matrix}{{Q_{1} = {\frac{\omega_{s}L_{1}}{R_{1}}1}},} & (18)\end{matrix}$

the magnitude impedance |Z_(1,0)| is approximately the sense coil's 302reactance ω_(s)L₁ so that equation (17) may be rewritten as

$\begin{matrix}{\frac{\Delta Z_{1}}{| Z_{1,0} |} \cong {\frac{\alpha_{13}^{2}Z_{3}^{*}}{\omega_{s}L_{1}}.}} & (19)\end{matrix}$

Substituting α₁₃ and Z₃ using equations (6), (7), (9), and (11),equation (19) may be expressed solely in terms of coupling factor k₁₃and the object's 110 Q-factor Q₃ as follows:

$\begin{matrix}{{\frac{\Delta Z_{1}}{| Z_{1,0} |} \cong {k_{13}^{2}\frac{Q_{3}}{1 + Q_{3}^{2}}( {1 - {jQ_{3}}} )}},} & (20)\end{matrix}$

For objects 110 with high enough Q-factor Q₃>>1, equation (20) may berewritten as

$\begin{matrix}{\frac{\Delta Z_{1}}{| Z_{1,0} |} \cong {k_{13}^{2}( {\frac{1}{Q_{3}} - j} )}} & (21)\end{matrix}$

Equation (21) shows that the imaginary part of the fractional changeΔZ₁/|Z_(1,0)| is approximately the squared coupling factor k₁₃ while thereal part is Q₃-times lower.

In some aspects of an object detection circuit 100, the fractionalchange |ΔZ₁|/|Z_(1,0)| may determine the required dynamic range of thevoltage measurement circuit 304. It may be appreciated that detection ofobjects 110 with a low fractional change |ΔZ₁|/|Z_(1,0)| requires a highdynamic range demanding components with higher performance in terms ofintrinsic thermal noise and quantization noise (e.g., if the voltagemeasurement circuit 304 includes an amplifier and/or ananalog-to-digital converter). Therefore, the fractional change|ΔZ₁|/|Z_(1,0)| may indirectly impact cost of an object detectioncircuit 100. Therefore, it may be desirable to increase a fractionalchange |ΔZ₁|/|Z_(1,0)| as produced by an object 110 at a given positionand orientation relative to the sense coil 302.

In another aspect, the fractional change |ΔZ₁|/|Z_(1,0)| may also atleast partially determine temperature sensitivity of the objectdetection circuit 100 as a consequence of a temperature dependentimpedance Z_(1,0) as previously discussed in connection with FIG. 3B.

In some aspects of an object detection circuit 100, it may be desirableto reduce a temperature sensitivity. Temperature sensitivity may bedefined e.g., with respect to the real part as the ratio of an impedancechange Re{δZ_(1,0)} due to a temperature change Δϑ to an impedancechange Re{ΔZ₁} produced by the object 110. This ratio e.g., with respectto the real part may be also expressed as the ratio of the fractionalchanges Re{δZ_(1,0)}/|Z_(1,0)| and Re{ΔZ}/|Z_(1,0)|

$\begin{matrix}{{\frac{{Re}\{ {\delta \; Z_{1,0}} \}}{{Re}\{ {\Delta Z_{1}} \}} = {\frac{{Re}\{ {\delta \; Z_{1,0}} \}}{| Z_{1,0} |} \cdot \frac{| Z_{1,0} |}{{Re}\{ {\Delta Z_{1}} \}}}},} & (22)\end{matrix}$

and for the imaginary part accordingly

$\begin{matrix}{\frac{{Im}\{ {\delta Z_{0}} \}}{{Im}\{ {\Delta Z_{1}} \}} = {\frac{{Im}\{ {\delta Z_{1,0}} \}}{Z_{1,0}} \cdot {\frac{Z_{1,0}}{{Im}\{ {\Delta Z_{1}} \}}.}}} & (23)\end{matrix}$

Equation (22) shows that temperature sensitivity with respect to thereal part may reduce if the fractional change Re{ΔZ₁}/|Z_(1,0)| isincreased, provided that the fractional change Re{(δZ_(1,0)}/|Z_(1,0)|does not increase likewise by this improvement. This may be also validfor the imaginary part as shown by Equation (23).

For an object 110 at a position close enough to the sense coil 302, thefractional change |ΔZ₁|/|Z_(1,0)| may be increased by reducing the sizeof the sense coil 302. In some implementations of the object detectioncircuit 100 of FIG. 1 based on a plurality of sense coils 106 a, 106 b,. . . , 106 n, a smaller sense coil size may be used which may result ina larger number of sense coils.

An increase of the fractional change |ΔZ₁|/|Z_(1,0)| for objects 110 atcertain positions and orientations may be also achieved with anoptimized design of sense coil 302. For example, a sense coil 302 may beimplemented with a spread (spiral) winding (as opposed to a moreconcentrated winding as illustrated in FIG.: 3A) which may provide alarger fractional change |ΔZ₁|/|Z_(1,0)| for objects 110 placed near thecenter of the sense coil 302.

In some other aspects of the object detection circuit 100 of FIG. 1,object detection sensitivity may be also defined with respect to noisee.g., in terms of the signal-to-noise ratio (SNR) as follows:

$\begin{matrix}{{{SNR} = {\frac{{{\Delta \; V_{1}}}^{2}}{V_{1,n}^{2}} = \frac{I_{1}^{2} \cdot {{\Delta \; Z_{1}}}^{2}}{V_{1,n}^{2}}}}.} & (24)\end{matrix}$

where V_(1,n) denotes an equivalent r.m.s. noise voltage at sensefrequency f_(s) in the bandwidth B_(m) of the voltage measurementcircuit 304. An equivalent noise voltage source V_(1,n) (not shown inFIG. 3C) may be considered in series to the impedance Z₁ in theequivalent circuit of FIG. 3C. This equivalent noise voltage source maygenerally include the effect of various circuit intrinsic and extrinsicnoise sources.

Circuit intrinsic noise may include thermal noise generated by theresistance R₁ and in the voltage measurement circuit 304. Noise producedin the voltage measurement circuit 304 may include thermal noise andquantization noise e.g., if the voltage measurement circuit 304 involvesan analog-to-digital converter (ADC). The circuit intrinsic noise mayalso include noise generated in the current source 306 e.g., a noisecurrent component producing a noise voltage across impedance Z₁. In someimplementations of a current source 306, this noise current may includeadditive noise, phase noise and amplitude modulated (AM) noise. Additivenoise may include thermal noise as generated in the circuit of thecurrent source 306. It may also include quantization noise if thecurrent source involves a digital signal synthesizes and adigital-to-analog converter (DAC). In some implementations, the effectof DAC quantization noise may appear as a deterministic error on thecurrent amplitude rather than a stochastic noise component. In someimplementations, the amplitude of current I₁ may be modulated by a lowfrequency noise component with a 1/f-characteristics, e.g., if thecurrent source 306 involves a DC supply voltage regulator that generateslow frequency noise. Therefore, some implementations of the objectdetection circuit 100 of FIG. 1 use a current source 306 and a voltagemeasurement circuit 304 optimized with respect to all sort of noisecomponents to maximize a sensitivity (e.g., the SNR as defined byEquation (24)) with respect to circuit intrinsic noise.

In some implementations of the object detection circuit 100 of FIG. 1using a sense element array 106 integrated together with the wirelesspower transfer structure 224 into housing 236 as shown in FIG. 2A.Switching noise in the wireless power electromagnetic field produced bythe power conversion circuit 222 may be inductively and/or capacitivelycoupled into sense coil 302 when the wireless power transfer system isactive. This circuit extrinsic noise component may include wideband andnarrowband noise (e.g., harmonics noise). Therefore, in someimplementations of a wireless power transfer system 200 including anobject detection circuit 100, the power conversion circuit 222 isoptimized with respect to noise e.g., by using extra filtering tosuppress noise in a sense frequency band. Moreover, the voltagemeasurement circuit 304 may be configured to minimize the effect ofcircuit extrinsic noise sources. In some implementations of an objectdetection circuit 100, the noise spectrum is monitored with a spectralresolution corresponding to the bandwidth B_(m) of the voltagemeasurement circuit 304 to identify potential sense frequencies f_(s)with lowest noise level and within some operational frequency limitsdetermined by the object detection circuit 100. As part of anoptimization, the object detection circuit 100 may select a sensefrequency f_(s) at a frequency providing lowest noise level. Suchprocedure may be used to maximize the SNR as defined by Equation (24))with respect to circuit extrinsic noise and thus to maximize sensitivityof the object detection circuit 100 e.g., in presence of switchingnoise.

FIG. 4A shows a circuit 400 illustrating another technique forinductively detecting a presence of an object 110. Circuit 400 uses acapacitor 420 having capacitance C to compensate for the reactanceω_(s)L₁ of the sense coil 302 having an inductance L₁. The circuit 400uses the same current source voltage measurement technique for asdescribed in connection with circuit 300 of FIG. 3A for measuring animpedance Z_(1c)=Z_(1c,0)+ΔZ₁ that now includes the reactance ofcompensation (tuning) capacitor 420. In some implementations of anobject detection circuit 100, capacitor 420 may not fully compensate forthe sense coil's 302 reactance at a specified sense frequency f_(s)(e.g., in the MHz range). However, the frequency of the current sourcedefining the sense frequency f_(s) may be adjustable so that theLC-circuit may be tuned for improved compensation (resonance) in theabsence of an object 110.

An equivalent circuit of the circuit 400 of FIG. 4A is illustrated inFIG. 4B. The sense coil 302 is represented by an equivalent inductanceL₁(ϑ) and an equivalent loss resistance R₁(ϑ), and the compensationcapacitor 420 by its equivalent capacitance C(ϑ), all three shown as afunction of temperature ϑ. As previously stated, inductance L₁(ϑ) andresistance R₁(ϑ) may include a change of an inductance and resistance,respectively, due to the presence of materials 310. Depending on theelectromagnetic properties of materials 310, this may be a decrease orincrease of the inductance L₁(ϑ) and resistance R₁(ϑ) as measured infree space. The object 110 is abstracted away by impedance change ΔZ₁ aspreviously discussed.

In case of perfect reactance compensation (tuning), the impedanceZ_(1c,0) as measured in absence of object 110 may only include theequivalent resistance R₁(ϑ). Assuming perfect compensation, thedetection sensitivity as defined by the fractional impedance change maybe expressed as

$\begin{matrix}{{\frac{\Delta Z_{1}}{Z_{{1c},0}} = \frac{\alpha_{13}^{2}Z_{3}^{*}}{R_{1}}},} & (25)\end{matrix}$

and in terms of the sense coil's 302 Q-factor Q₁, coupling factor k₁₃,and the object's 110 Q-factor Q₃ as

$\begin{matrix}{{\frac{\Delta Z_{1}}{Z_{{1c},0}} \cong {k_{13}^{2}\frac{Q_{1}Q_{3}}{1 + Q_{3}^{2}}( {1 - {jQ_{3}}} )}},} & (26)\end{matrix}$

For non-ferromagnetic objects 110 with high enough Q-factor Q₃>>1,equation (26) may be rewritten as

$\begin{matrix}{{\frac{\Delta Z_{1}}{Z_{{1c},0}} \cong {k_{13}^{2}{Q_{1}( {\frac{1}{Q_{3}} - j} )}}}.} & (27)\end{matrix}$

showing that the fractional change |ΔZ₁|/|Z_(1c,0)| that may beachievable with the circuit 400 using reactance compensation is Q₁-timeshigher than the fractional change |ΔZ₁|/|Z_(1,0)| obtained with thecircuit 300 of FIG. 3A. Therefore, in some aspects of an objectdetection circuit 100, reactance compensation as illustrated by circuit400 of FIG. 4A may increase sensitivity for various objects.

In some implementations of the object detection circuit 100 of FIG. 1based on capacitively compensated impedance sensing, impedancemeasurement may be subjected to measurement errors. Reactancecompensation may provide a mean for accurate calibration of theimpedance measurement e.g., with respect to the angle arg{Z_(12c)} forpurposes as previously discussed in connection with the circuit 300 ofFIG. 3A. In some exemplary implementations of an object detectioncircuit 100, the frequency f_(s) of the current source 306 is tuned suchthat the magnitude impedance |Z_(1c,0)| becomes substantially a minimumin absence of the object 110, meaning that |Z_(1c,0)|≅R₁. Knowing thatin absence of the object 110, the minimum |Z_(1c,0)| ideally correspondsto a zero angle (arg{Z_(1c,0)}=0), the object detection circuit 100 maycorrect as part of a calibration procedure the actually measuredimpedance measurement by rotating the impedance plane so that theIm{Z_(1c,0)} vanishes.

In certain implementations of an object detection circuit 100, reactancecompensation may allow the dynamic range requirements of the voltagemeasurement circuit 304 to be reduced by a factor corresponding to theQ-factor Q₁ of the sense coil 302, neglecting any margin for temperaturedrift. For example, assuming a Q-factor of Q₁=40, the required dynamicrange may reduce by 32 dB. This may result in a general increase of therobustness of the voltage measurement circuit 304 against effects suchas temperature drift, moisture, thermal and quantization noise, noiseand instability on DC power supplies, and eventually in reduced circuitcomplexity and cost savings in some components (e.g., low noiseamplifier, analog-to-digital converter, power supplies).

In some implementations of the object detection circuit 100 of FIG. 1using a voltage measurement circuit 304 with an input voltagelimitation, reactance compensation may also allow a sense current I₁ tobe increased by a factor of Q₁, which may in turn result in an increaseof the signal-to-noise ratio (SNR) with respect to external noise (e.g.,high frequency noise and harmonics in the electromagnetic fields asgenerated for the wireless power transfer) that is coupled into thesense coil 302.

Moreover, in some implementations of an object detection circuit 100,reactance compensation using a series capacitor 420 together with avoltage measurement circuit 304 that presents a high input impedance atsense frequency f_(s) and a low impedance at low frequencies may form ahigh pass filter to attenuate low frequency signal components e.g., atthe wireless power transfer frequency f_(wpt) (e.g., 85 kHz) asdescribed in more detail in connection with FIG. 10. Note thatrelatively strong low frequency components may be induced into the sensecoil 302 during wireless power transfer. This may result in relaxedrequirements for the current source 306 and the voltage measurementcircuit 304 with respect to the dynamic range, overvoltage capability,etc.

However, in certain aspects reactance compensation may not reduce atemperature sensitivity of the object detection circuit 100 as it may bedefined in analogy to equations (22) and (23) respectively, sincereactance compensation may not reduce a change δZ_(1c,0) due to a changein temperature Δϑ. In contrast, the change of Im{δZ_(1c,0)} may begenerally larger than Im{δZ_(1,0)} if compensation capacitor 420 is alsoa function of temperature ϑ.

Therefore, in some implementations of an object detection circuit 100,temperature sensitivity is reduced by either using a capacitor with alow temperature coefficient (e.g., NP0-type) or with a temperaturecoefficient that compensates or partially compensates for thetemperature drift of the sense coil's impedance Z_(1,0).

While circuits 300 and 400 of FIGS. 3A and 4A illustrate a technique fordetecting an object 110 based on measuring a self-impedance Z₁ or acapacitively compensated self-impedance Z₁, of a sense coil 302,respectively, the circuit 500 of FIG. 5A illustrates another techniquefor detecting an object 110 based on measuring a transimpedance Z₁₂between a first (primary) sense coil 512 with inductance L₁ and a second(secondary) sense coil 514 with inductance L₂ of a sense coilarrangement 510. As for the self-impedance technique, a change ΔZ₁₂ ofthe transimpedance Z₁₂ relative to the transimpedance Z_(12,0) in theabsence of the object 110 may indicate a presence of the object 110 inproximity of at least one of the two sense coils 512 and 514.

In some implementations, primary sense coil 512 and secondary sense coil514 of the sense coil arrangement 510 are planar multi-turn coils andare disposed such that there is magnetic coupling between the two sensecoils (e.g., sense coils may be overlapping as shown in FIG. 5A). Achange of transimpedance ΔZ₁₂ may be also experienced when the two sensecoils 512 and 514 are integrated together with the wireless powertransfer structure 224 into housing 236 as shown in FIG. 2A due topresence of materials 310 as previously discussed. Presence of materials310 is indicated in FIG. 5A by the shaded area. Materials 310 may alsoinclude a dielectric substrate that carries the two sense coils 512 and516 or a dielectric medium the sense coils are embedded in (e.g., incase of a printed circuit board (PCB) design). The effects of materials310 may generally produce a change in the sense coils' 512 and 514self-impedance Z_(1,0) and Z_(2,0), respectively, and in thetransimpedance Z_(12,0) relative to these impedances as measured in freespace. The effects of materials 310 are considered already included inZ_(1,0), Z_(2,0) and Z_(12,0).

The circuit 500 of FIG. 5A illustrates a technique for measuring atransimpedance Z₁₂ based on the current source voltage measurementapproach. A sinusoidal current I₁ with a defined frequency, amplitude,and phase provided by a current source 306 is applied to the primarysense coil 512 and the open-circuit voltage V₂ is measured across theterminals of the secondary sense coil 514. The open-circuit voltage V₁is measured using a sensitive high impedance voltage measurement circuit304 so that there is virtually zero current at the terminals of thesecondary sense coil 514 (I₂≅0). The current source 306 and the voltagemeasurement circuit 304 may be part of the measurement and detectioncircuit 108 with reference to FIG. 1. The voltage measurement circuit304 may be frequency selective (narrowband) tuned to the sense frequencyf_(s) (frequency of the current source) as previously discussed inconnection with the circuit 300 of FIG. 3A. The transimpedanceZ₁₂=Z_(12,0)+ΔZ₁₂ between the two sense coils 512 and 514 may bedetermined by dividing the measured voltage V_(2,0)+ΔV₂ by the defined(known) current I₁, which may be expressed more formally as

$\begin{matrix}{{Z_{12} = {{Z_{{12},0} + {\Delta Z_{12}}} = \frac{V_{2,0} + {\Delta V_{2}}}{I_{1}}}},} & (28)\end{matrix}$

Using an ideal current source 306, a change ΔZ₁₂ in the transimpedanceZ₁₂ e.g., due to presence of the object 110 manifests in a change ΔV₂ inthe voltage V₂ while the current I₁ remains unaffected. Therefore,measuring a change ΔV₂ in the voltage V₂ may be equivalent to measuringa change ΔZ₁₂ in the transimpedance Z₁₂. In other words, the voltage V₂may be indicative of the transimpedance Z₁₂. The current source 306 andthe voltage measurement circuit 304 used in an implementation of thecircuit 500 of FIG. 5A may be characterized by a quasi-ideal currentsource and a quasi-ideal voltage measurement circuit, respectively, asdefined in connection with the circuit 300 of FIG. 3A.

As opposed to the self-impedance as measured at a single sense coil 302,the transimpedance as measured between a pair of sense coils (e.g., 512and 514) depends on the measurement technique (e.g., on loadingconditions of sense coils 512 and 514). The transimpedance as measuredwith the current source voltage measurement approach illustrated by thecircuit 500 of FIG. 5A may be also referred to as the mutual impedance.Other measurement techniques (not illustrated herein) e.g., by applyinga sinusoidal voltage V₁ with a defined frequency, amplitude, and phaseprovided by a voltage source to the primary sense coil 512 and measuringthe short circuit current I₂=I_(2,0)+ΔI₂ at the secondary sense coil 514using a sensitive (e.g., frequency selective) low impedance currentmeasurement circuit generally measure a different transimpedance asfurther discussed below in connection with FIG. 5C.

An equivalent circuit of the circuit 500 of FIG. 5A including a circuitmodel of the object 110 is shown in FIG. 5B. In this equivalent circuit,the primary sense coil 512 and secondary sense coil 514 are eachrepresented by an equivalent inductance L₁(ϑ) and L₂(ϑ), respectively,and by an equivalent loss resistance R₁(ϑ) and R₂(ϑ), all shown as afunction of temperature ϑ. The equivalent loss resistances R₁(ϑ) andR₂(ϑ) may each include a first resistance component due to losses in thesense coil's conductive structure (e.g., copper wires or PCB traces) anda second resistance component due to loss effects in materials 310 aspreviously discussed in connection with FIG. 3B. Magnetic couplingbetween sense coils 512 and 514 is represented by a complex couplingfactor k ₁₂(ϑ), also shown as a function of temperature ϑ as furtherexplained below. As previously stated, the equivalent inductances andresistance L₁(ϑ), L₂(ϑ), R₁(ϑ) and R₂(ϑ), respectively, as well as thecoupling factor k ₁₂(ϑ) may include a change due to the presence ofmaterials 310. Since the electromagnetic properties of materials 310 maygenerally be temperature dependent, some temperature dependence may beexpected for the coupling factor k ₁₂ (ϑ). The change in k ₁₂ (ϑ) due tomaterials 310 may include a real and imaginary component due to reactiveand resistive (loss) effects, respectively, in materials 310 but alsodue to mutual loss effects e.g., an eddy current loss effect in thesecondary sense coil 514 produced by the magnetic field of the primarysense coil 512. The real component relates to the mutual inductance M₁₂expressed as

M ₁₂(ϑ)=Re{ k ₁₂(ϑ)}√{square root over (L ₁(ϑ)L ₂(ϑ))},  (29)

while the imaginary component may be considered related to a mutualresistance expressed as

R ₁₂(ϑ)=ω_(s) Im{ k ₂(ϑ)}L ₁(ϑ)L ₂(ϑ).  (30)

FIG. 5B shows each of the primary sense coil 512 and the secondary sensecoil 514 magnetically coupled with coupling factor k₁₃ and k₂₃,respectively, to the object 110 represented by an LR-Model (L₃, R₃) aspreviously discussed e.g., with reference to FIG. 3B. The effect ofpresence of the object 110 in proximity of coil arrangement 510 may bethreefold. First, it produces a change ΔZ₁ in the impedance Z₁ as itwould be measured at the terminals of sense coil 512. Second, itproduces a change ΔZ₂ in the impedance Z₂ as it would be measured at theterminals of sense coil 514. Third, it produces a change ΔZ₁₂ in themutual impedance Z₁₂ as measured between sense coil 512 and sense coil514. Equations for these changes may be obtained from circuit analysisof the equivalent circuit model depicted in FIG. 5B. Defining theimpedance of the primary sense coil 512 in absence of the object 110

Z _(1,0) =R ₁ +jω _(s) L ₁,  (31)

the impedance of the secondary sense coil 514 in absence of the object110

Z _(2,0) =R ₂ +jω _(s) L ₂,  (32)

the impedance of the object 110

Z ₃ =R ₃ +jω _(s) L ₃,  (33)

the mutual impedance between sense coil 512 and 514 in absence of theobject 514

Z _(12,0) =R ₁₂ +jω _(s) M ₁₂=ω_(s) k ₁₂√{square root over (L ₁ L₂)},  (34)

the mutual impedance between primary sense coil 512 and the model of theobject 110

Z ₁₃ =jω _(s) M ₁₃=ω_(s) k ₁₃√{square root over (L ₁ L ₃)},  (35)

the mutual impedance between secondary sense coil 514 and the model ofthe object 110

Z ₂₃ =jω _(s) M ₂₃ =ωs k ₂₃√{square root over (L ₂ L ₃)},  (36)

and applying Kirchhoff's current and voltage laws provides the followingsystem of equations:

Z _(1,0) I ₁ −Z ₁₂ I ₂ −Z ₁₃ I ₃ −V ₁=0,  (37)

Z ₁₂ I ₁ −Z _(2,0) I ₂ −Z ₂₃ I ₃ −V ₂=0,  (38)

Z ₁₃ I ₁ −Z ₂₃ I ₂ −Z ₃ I ₃=0.  (39)

Equations for the changes in Z₁, Z₂, and Z₁₂ may be obtained byeliminating the variable I₃ in equations (37) and (38) by substituting

$\begin{matrix}{I_{3} = {{\frac{Z_{13}}{Z_{3}}I_{1}} - {\frac{Z_{23}}{Z_{3}}I_{2}}}} & (40)\end{matrix}$

yielding the following equation pair

$\begin{matrix}{{{( {Z_{1,0} - \frac{Z_{13}^{2}}{Z_{3}}} )I_{1}} - {( {Z_{{12},0} - \frac{Z_{13}Z_{23}}{Z_{3}}} )I_{2}} - V_{1}} = 0} & (41) \\{{{( {Z_{{12},0} - \frac{Z_{13}Z_{23}}{Z_{3}}} )I_{1}} - {( {Z_{2,0} - \frac{Z_{23}^{2}}{Z_{3}}} )I_{2}} - V_{2}} = 0} & (42)\end{matrix}$

The second term of the expressions in brackets may be considered thechange in impedance produced by the object 110 which may now beexpressed as

$\begin{matrix}{{\Delta Z_{1}} = {{- \frac{Z_{13}^{2}}{Z_{3}}} = {{{- \frac{Z_{13}^{2}}{{Z_{3}}^{2}}}Z_{3}^{*}} = {\alpha_{13}^{2}Z_{3}^{*}}}}} & (43) \\{{\Delta Z_{2}} = {{- \frac{Z_{23}^{2}}{Z_{3}}} = {{{- \frac{Z_{23}^{2}}{{Z_{3}}^{2}}}Z_{3}^{*}} = {\alpha_{23}^{2}Z_{3}^{*}}}}} & (44) \\{{\Delta Z_{12}} = {{- \frac{Z_{13}Z_{23}}{Z_{3}}} = {{{- \frac{Z_{13}Z_{23}}{{Z_{3}}^{2}}}Z_{3}^{*}} = {\alpha_{13}\alpha_{23}Z_{3}^{*}}}}} & (45)\end{matrix}$

using the scalar transformation factors

$\begin{matrix}{\alpha_{13} = \frac{\omega_{s}M_{13}}{Z_{3}}} & (46) \\{\alpha_{23} = \frac{\omega_{s}M_{23}}{Z_{3}}} & (47)\end{matrix}$

Equations (40), (41), and (45) show that each of the three impedancechanges are proportional to Z₃* thus reflecting the characteristics ofthe object 110 in terms of its Q-factor Q₃ as previously discussed inconnection with FIG. 3B.

FIG. 5C shows an equivalent circuit of the circuit 500 of FIG. 5A. Thesense coil arrangement 512 is represented by a ‘T’-equivalent circuitmodel (transformer model) comprised of a first series branch impedance

Z ₁ −Z ₁₂ =R ₁ −R ₁₂ +jω _(s)(L ₁ −M ₁₂)+ΔZ ₁ −ΔZ ₁₂,  (48)

a second series branch impedance

Z ₂ −Z ₁₂ =R ₂ −R ₁₂ +jωs(L ₂ −M ₁₂)+ΔZ ₂ −ΔZ ₁₂,  (49)

and a shunt branch impedance

Z ₁₂ =R ₁₂ +jω _(s) M ₁₂ +ΔZ ₁₂.  (50)

Each of the three branch impedances is composed of an inductance,resistance and an impedance change due to presence of the object 110.More precisely, the equivalent circuit of FIG. 5B shows the model of theobject 110 abstracted away by impedance change ΔZ₁−ΔZ₁₂ and ΔZ₂−ΔZ₁₂ inthe first and second series branch impedance Z₁−Z₁₂ and Z₂−Z₁₂,respectively, and by a change of mutual impedance ΔZ₁₂ in the shuntbranch impedance Z₁₂. Though not shown in FIG. 5C for illustrativepurposes and simplicity, each of the equivalent circuit elements may betemperature dependent.

Given the current source voltage measurement approach dictating acurrent I₁ in the primary sense coil 512 and measuring the open-circuitvoltage V₂ (I₂≅0) at the terminals of the secondary sense coil 514, itmay be appreciated that leakage inductances L₁−M₁₂ and L₂−M₁₂,equivalent series resistances R₁−R₁₂ and R₂−R₁₂, and impedance changesΔZ₁−ΔZ₁₂ and ΔZ₂−ΔZ₁₂ have ideally no effect in the transimpedancemeasurement. Transimpedance as measured with the current source voltagemeasurement approach ideally is determined by the shunt branch impedancethat is referred to as the mutual impedance

$\begin{matrix}{{Z_{12} = {{Z_{{12},0} + {\Delta Z_{12}}} = {{R_{12} + {j\omega_{s}M_{12}} + {\Delta Z_{12}}} = \frac{V_{2,0} + {\Delta V_{2}}}{I_{1}}}}}.} & (51)\end{matrix}$

Analogously to equation (16), the sensitivity of the object detectioncircuit 100 of FIG. 1 based on the mutual impedance sensing techniquemay be defined as the mutual impedance change ΔZ₁₂ as produced inpresence of an object 110 normalized to |Z_(12,0)| that is the magnitudeof the mutual impedance as measured in absence of the object 110. Thisnormalized mutual impedance change (sensitivity) may be also referred toas the fractional change ΔZ₁₂/|Z_(1,0)|. Using equation (45), thefractional change due to presence of a non-ferromagnetic object 110 maybe expressed as

$\begin{matrix}{{\frac{\Delta Z_{12}}{Z_{12,0}} = \frac{\alpha_{13}\alpha_{23}Z_{3}^{*}}{Z_{12,0}}}.} & (52)\end{matrix}$

For a sense coil arrangement 510 with high enough mutual inductance

ω_(s) M ₁₂ >>R ₁,  (53)

the magnitude mutual impedance |Z_(12,0)| is approximately the mutualreactance ω_(s)M₁₂ so that equation (52) may be rewritten as

$\begin{matrix}{\frac{\Delta Z_{12}}{Z_{12,0}} \cong {\frac{\alpha_{13}\alpha_{23}Z_{3}^{*}}{{R_{12} + {j\; \omega_{s}M_{12}}}}.}} & (54)\end{matrix}$

Substituting α₁₃, α₂₃ and Z₃ using equations (46), (47), (9), and (11),equation (54) may be expressed analogously to equation (20) solely interms of coupling factors k₁₃, k₂₃, k ₁₂ and the object's 110 Q-factorQ₃ as follows:

$\begin{matrix}{{\frac{\Delta Z_{12}}{Z_{12,0}} \cong {\frac{k_{13}k_{23}}{{\underset{\_}{k}}_{12}}\frac{Q_{3}}{1 + Q_{3}^{2}}( {1 - {jQ_{3}}} )}},} & (55)\end{matrix}$

For objects 110 with high enough Q-factor Q₃>>1, equation (55) may berewritten as

$\begin{matrix}{\frac{\Delta Z_{12}}{Z_{12,0}} \cong {\frac{k_{13}k_{23}}{{\underset{\_}{k}}_{12}}( {\frac{1}{Q_{3}} - j} )}} & (56)\end{matrix}$

For the special case of a tightly coupled sense coil arrangement 510with |k ₁₂|≅<1 (e.g., using two identical sense coils on top of eachother and with zero displacement as shown by FIG. 5F, it may be aconsequence that both sense coils 512 and 514 provide about equalcoupling to the object 110 (k₁₃≅k₂₃). In this special case, equation(56) may be rewritten as

$\begin{matrix}{\frac{\Delta \; Z_{12}}{Z_{12,0}} \cong {k_{13}^{2}( {\frac{1}{Q_{3}} - j} )}} & (57)\end{matrix}$

Equation (57) indicates that the mutual impedance sensing technique hasthe potential to provide a sensitivity (fractional change) equal to thatof self-impedance sensing (Equation (21)) assuming equal coupling to theobject 110 for both sensing methods (circuit 300 of FIG. 3A and circuit500 of FIG. 5A).

Potentially equal sensitivity may be also expected with respect to noiseas previously discussed in connection with FIG. 3C. An equal equivalentnoise voltage level (V_(1,n)=V_(2,n)) may be assumed for both circuit300 and 500 if circuit extrinsic noise is the predominant noise sourceand if sense coils 302 and 514 are identical. Furthermore, assuming anequal current level I₁ and approximately equal coupling to the object110 in both circuits 300 and 500, and for primary 512 and secondarysense coil 514 (k₁₃≅k₂₃), equal SNR may be expected from both sensingmethods (circuits 300 and 500)

$\begin{matrix}{{SNR} = {\frac{I_{1}^{2} \cdot {{\Delta \; Z_{1}}}^{2}}{V_{1,n}^{2}} \cong {\frac{I_{1}^{2} \cdot {{\Delta \; Z_{12}}}^{2}}{V_{2,n}^{2}}.}}} & (58)\end{matrix}$

However, in case k₁₃≠k₂₃ (e.g., primary and secondary sense coil arephysically displaced e.g., as illustrated by FIG. 5D), a lower SNR maybe expected from mutual impedance sensing (circuit 500).

As previously stated, the change in mutual impedance ΔZ₁₂ is angle-true(arg{ΔZ₁}=−arg{Z₃}), meaning that it reflects the electromagneticcharacteristics (impedance Z₃) of the object 110. This feature may allowthe object detection circuit 100 to discriminate certain categories ofobjects and/or increase a sensitivity for certain categories of objectsas previously discussed in connection with the self-impedance sensingtechnique (circuit 300) as illustrated by FIG. 3A.

In some aspect, other transimpedance measurement techniques may be usede.g., the voltage source current measurement approach as previouslydescribed in connection with the circuit 300 of FIG. 3A. An expressionfor the change ΔZ₁₂ in transimpedance Z₁₂ as resulting with the voltagesource current measurement approach may be also obtained from the systemof equations (41) and (42). Setting V₂=0 (the voltage across the currentmeasurement circuit) and solving for the current I₂ yields for thechange in transimpedance due to proximity of the object 110

$\begin{matrix}{{\Delta \; Z_{12}} = {( {{\frac{Z_{1,0}}{Z_{12,0}}\alpha_{13}^{2}} + {\frac{Z_{2,0}}{Z_{12,0}}\alpha_{23}^{2}} - {\alpha_{13}\alpha_{23}}} ){Z_{3}^{*}.}}} & (59)\end{matrix}$

It may be appreciated that the expression in brackets generally is anon-scalar (complex) transformation factor. Therefore, the change ΔZ₁₂in transimpedance may be no more proportional to the object's 110sign-inverted impedance −Z₃, hence arg{ΔZ₁₂}≠−arg{Z₃}). Moreover, theresulting angle arg{ΔZ₁₂} may vary with position and orientation of theobject 110 relative to the sense coil arrangement 510. In some aspectsof transimpedance sensing, this finding may be considered as adifficulty of the voltage source current measurement approach.

In some implementations of the object detection circuit 100 of FIG. 1based on the mutual impedance sensing technique, the object detectioncircuit 100 is configured to calibrate the mutual impedance measurementsfor purposes as previously described in connection with FIGS. 3A to 3D.However, implementation of the calibration may result in higher circuitcomplexity and cost in certain aspects, e.g., if compared to aspects ofcapacitively compensated self-impedance sensing using sense frequencytuning as discussed in connection with FIG. 4A.

FIGS. 5D to 5H are cut views illustrating various exemplaryimplementations of planar sense coil arrangements 510 as they may beused for transimpedance (e.g., mutual impedance) sensing.

FIG. 5D illustrates an implementation of a sense coil arrangement 510where sense coils 512 and 514 are coplanar and adjacent. This sense coilarrangement 510 may apply to an implementation using a plurality ofsubstantially equal sense elements 106 a, 106 b, . . . , 106 n eachincluding a single planar sense coil arranged in an array 106 withreference to FIG. 2A. Pairs of neighboring sense coils (e.g., sense coil106 a and 106 b) may be temporarily configured using a multiplexercircuitry as further discussed with reference to FIG. 10. A first pairmay be configured by sense coils 106 a and 106 b, a second pair may beconfigured by sense coils 106 b and 106 c, a third pair may beconfigured by sense coils 106 c and 106 d etc. Sense coils pairs thatare temporarily (e.g., sequentially) configured may be overlapping. Itmay be also appreciated that in such implementation or operation, thenumber of double sense coil arrangements 510 that can be potentiallyconfigured is larger than the number N of sense elements of the array106.

FIG. 5E illustrates another implementation of a sense coil arrangement510 where sense coils 512 and 514 are partially overlapping. This sensecoil arrangement 510 may apply to an implementation using a plurality ofsubstantially equal sense elements 106 a, 106 b, . . . , 106 n eachincluding a single planar sense coil arranged in an array 106 having afirst and second plane. Sense coils in the first plane are offsetrelative to the sense coils in the second plane by half of the width ofthe sense coil. Pairs of overlapping sense coils e.g., with a primarysense coil 512 in a first plane and a secondary sense coil 514 in asecond plane may be temporally (e.g., sequentially) configured using amultiplexer circuitry as further discussed with reference to FIG. 10. Afirst pair may be configured by sense coil 106 a in the first plane andsense coil 106 b in the second plane, a second pair may be configured bysense coil 106 b in the second plane and sense coil 106 c in the firstplane, a third pair may be configured by sense coils 106 c in the firstplane and sense coil 106 d in the second plane, etc. Sense coil pairstemporarily configured in this way may be overlapping.

FIG. 5F illustrates a further implementation of a sense coil arrangement510 where sense coils 512 and 514 are fully overlapping (on top of eachother). This sense coil arrangement 510 may apply to an implementationusing a plurality of sense elements 106 a, 106 b, . . . , 106 n arrangedin an array 106. Each of the plurality sense elements 106 a, 106 b, . .. , 106 n includes a pair of planar sense coils stacked on top of eachother. This sense coil arrangement 510 may be also considered as abifilar winding structure with a first winding disposed in a first planeand a second winding disposed in a second plane.

FIG. 5G illustrates yet another exemplary implementation of a sense coilarrangement 510 where sense coils 512 and 514 are coplanar and arrangedinside of each other. This sense coil arrangement 510 may apply to animplementation using a plurality of sense elements 106 a, 106 b, . . . ,106 n arranged in an array 106. Each sense element includes a pair ofcoplanar sense coils arranged inside each other.

FIG. 5H illustrates yet a further implementation of a sense coilarrangement 510 where sense coils 512 and 514 are coplanar andinterleaved. This sense coil arrangement 510 may apply to animplementation using a plurality of sense elements 106 a, 106 b, . . . ,106 n arranged in an array 106 with reference to FIG. 2A. Each senseelement includes a pair of interleaved coplanar sense coils. This sensecoil arrangement 510 may be also considered as a bifilar windingstructure with a first and a second winding in the same plane.

FIG. 6A shows a circuit 600 to illustrate another technique based on themutual impedance sensing approach as previously described with referenceto FIGS. 5A to 5H. As in the circuit 500 of FIG. 5A, the mutualimpedance Z₁₂=Z_(12,0)+ΔZ₁₂ in presence of the object 110 is measuredusing the current source voltage measurement technique and a sense coilarrangement 510 (e.g., sense coil arrangement 510 of FIG. 5E) includingprimary sense coil 512 with inductance L₁ and secondary sense coil 514with inductance L₂. FIG. 6A shows the sense coil 514 displaced relativeto the sense coil 512 by an amount D. FIG. 6A also indicates a firstmagnetic flux Φ₁ through the overlap area and a second flux Φ₂ in theopposite direction through the non-overlap area of sense coil 514. Theremay be a displacement D where the net magnetic flux ΔΦ=Φ₁−Φ₂ generatedby the primary sense coil 512 and passing through the secondary sensecoil 514 is virtually zero (flux balance) resulting in virtually zerocoupling (|k ₁₂|≅0) in absence of the object 110. However, in presenceof materials 310 as previously described in connection with FIG. 3A,perfect flux balance (zero coupling) may not be achievable. Losses inmaterials 310 may cause a small out-of-phase component in the flux Φ₂relative to flux Φ₁ so that the net flux ΔΦ may never vanish entirelyfor any displacement D within some limits. In an aspect, thedisplacement D between the primary sense coil 512 and secondary sensecoil 514 may be adjusted to reduce coupling. In some implementations,the level of coupling may be negligible or at least reduced to below athreshold. In some aspects of an object detection circuit 100, thedisplacement D may be adjusted by design for minimum magnitude coupling|k ₁₂| between sense coil 512 and 514 corresponding to minimum magnitudemutual impedance |Z_(12,0)|=|R₁₂+jω_(s)M₁₂| in absence of the object110. A mutual impedance sensing technique operating at or near fluxbalance may be referred to as flux balanced mutual impedance sensing.

In some implementations of an object detection circuit 100, fluxbalanced mutual impedance sensing is used to increase a sensitivity(fractional change ΔZ₁₂/|Z_(12,0)|) as defined by Equation (54) formutual impedance sensing. It may be appreciated that the maximumsensitivity as achievable with the circuit 600 of FIG. 6A by designingthe sense coil arrangement 510 with an optimum displacement D providingminimum magnitude coupling |k ₁₂|=|k ₁₂≡_(min) results in asubstantially higher sensitivity compared to the sensitivity of thecircuit 500 of FIG. 5A e.g., designed for maximum coupling |k ₁₂|≅<1.The sensitivity gain in this example may be quantified approximately by1/|k ₁₂|_(min).

FIG. 6B shows a simplified equivalent circuit of the circuit 600 of FIG.6A. The sense coil arrangement 512 is represented by a simplified ‘T’equivalent circuit model neglecting first and second series branchimpedance as previously described with reference to FIG. 5C. Thissimplification may be permissible for the current source voltagemeasurement approach, since first and second series branch impedancehave virtually no impact in the mutual impedance measurement. Thissimplified equivalent circuit solely comprises the shunt branchimpedance Z₁₂ including the residual mutual inductance M₁₂, a residualresistance R₁₂, and mutual impedance change ΔZ₁₂ that represents theobject 110 abstracted away.

Flux balanced impedance sensing may be also implemented using a specialsense coil arrangement 510 as illustrated by circuit 600 in FIG. 6C. Inthis exemplary implementation, the primary sense coil 512 is of adifferent topology than the secondary sense coil 512 such that the netmagnetic flux ΔΦ=Φ₁−Φ₂ through the secondary sense coil 514 and thuscoupling |k ₁₂| is virtually zero (or close thereto) in absence of theobject 110. FIG. 6C shows the primary sense coil 512 as a “figure eight”or “DD”-type coil while the secondary sense coil 514 is shown as a“circular”-type coil. It may be appreciated that there exist otherimplementations or combinations providing flux balance in the secondarysense coil 514. In some example implementations, the primary sense coil512 is a “circular”-type coil and the secondary 514 is a “DD”-type coil.

In some implementations of the object detection circuit 100 of FIG. 1based on the flux balanced mutual impedance sensing technique, the senseelement array 106 is integrated together with the wireless powertransfer structure 224 into housing 236 as illustrated by FIG. 2B. Forsuch implementations, the design of the sense elements of the pluralityof sense elements 106 a, 106 b, . . . , 106 n (sense element array 106)may require some modifications to substantially compensate for the fluxunbalancing effects of materials 310 as previously defined in connectionwith FIG. 3A. These design modifications may be individual for eachsense element of the plurality of sense elements 106 a, 106 b, . . . ,106 n. However, even with such modifications, the minimum achievablecoupling |k ₁₂|_(min) may not be achieved in each of the plurality ofsense elements e.g., because the integration impact is difficult topredict or may vary over time due to mechanical changes, ageing,temperature variations, etc. resulting in some loss of sensitivity.Therefore, adjusting flux balance in situ may be desirable.

Regardless of the sense coil arrangement 510 used for flux balancedmutual impedance sensing, there may exist positions of the object 110where flux remains virtually balanced even for an object 110 in closeproximity to the sense coil arrangement 510, meaning reduced orvirtually zero sensitivity (ΔZ₁₂≅0). However, it shall be noticed thatthe blind spot (sum of all positions where sensitivity is below athreshold) may be extremely narrow. Therefore, the likelihood that anobject 110 comes to a rest in a blind spot may be very low. Thesepositions may also somewhat dependent on type, size, and orientation ofthe object 110. In some implementations based on the sense coilarrangement 510 of FIG. 6C, the blind spot e.g., for a coin may be foundin a narrow area along its vertical symmetry axis (not shown in FIG. 6C)if the coin rests on one of its large surfaces in a plane in parallel tothe planar sense coil arrangement 510 and if the center of the coindefines the coin's position. In some implementations using a sense coilarray (e.g., sense coil array 106), blind spots are reduced oreliminated using overlapping sense coil arrangements 510.

FIG. 7A shows a circuit 700 illustrating a further technique which maybe also referred to as flux balanced mutual impedance sensing. Thecircuit 700 is also based on the current source voltage measurementapproach. It includes a sense coil arrangement 710 of two primary sensecoils 512 with inductance L₁ and 516 with inductance L₃ and a secondarysense coil 514 with inductance L₂. Each of the primary sense coils 512and 516 is driven by a first and second current source 306 and 308delivering currents I₁ and I₂, respectively, e.g., with a sinusoidalcurrent in the MHz range. The secondary sense coil 514 is electricallyconnected to the voltage measurement circuit 504 for measuring theopen-circuit output voltage V₃. The second current source 308 deliveringcurrent I₂ is controllable in amplitude and phase so that the circuit700 may be operated such that the magnetic flux components as generatedby sense coil 512 and 516 and passing through the secondary sense coil514 cancel out (substantially zero flux or at least very low flux)resulting in substantially zero output voltage (V₃≅0) in absence of theobject 110. FIG. 7 also shows an object 110 at an exemplary position inproximity of the sense coil arrangement 710. Since the output voltage V₃is a function of two primary currents I₁ and I₂, the definition of atransimpedance (e.g., Equation (28)) may generally not apply. Therefore,this technique may rely on measuring the output voltage V₃ and theobject 110 is detected based on a change ΔV₃ in the output voltage V₃relative to the voltage V_(3,0) as measured in absence of the object110.

FIG. 7B shows an equivalent circuit of the circuit 700 includingequivalent loss resistances R₁, R₂, and R₃ of sense coils 512, 516, and514, respectively. Coupling between the first primary sense coil 512 andsecondary sense coil 514 and between the second primary sense coil 516and secondary sense coil 514 is represented by complex coupling factorsk ₁₃ and k ₂₃, respectively, as previously discussed in connection withFIG. 3B. Coupling between the two primary sense coils 512 and 516 mayalso exist but is omitted in FIG. 7B as it generally does not matter forthe current source voltage measurement approach. It also includes anequivalent circuit model of the object 110 (L₄, R₄) and correspondingcoupling to each sense coil (512, 514, and 516) represented by couplingfactors k₁₄, k₃₄, and k₂₄, respectively.

FIG. 7C shows a simplified equivalent circuit of the circuit 700 toillustrate the voltage cancellation effect in output voltage V₃. Thisequivalent circuit includes a simplified ‘T’-equivalent circuit(transformer) models reduced to the shunt branch (mutual) impedances Z₁₃and Z₂₃, respectively, for each of the two transmission paths I₁ to V₃and I₂ to V₃. Each shunt branch impedance Z₁₃ and Z₂₃ includes a mutualinductance M₁₃ and M₂₃, an equivalent loss resistance R₁₂ and R₂₃corresponding to the imaginary part of coupling factor k ₁₃ and k ₂₃,respectively. Each of the shunt branch impedances Z₁₃ and Z₂₃ alsoincludes a mutual impedance change ΔZ₁₃ and ΔZ₂₃ representing the modelof the object 110 abstracted away. FIG. 7C also indicates the twoprimary currents I₁ and I₂ and the resulting partial voltages V₁₃ andV₂₃ across shunt branch impedance Z₁₃ and Z₂₃, respectively. Using abovedefinitions, the output voltage V₃=V_(3,0)+ΔV₃ may be expressed as

V ₃ =V _(3,0) +ΔV ₃=(Z ₁₃ +ΔZ ₁₃)I ₁+(Z ₂₃ +ΔZ ₂₃)I ₂,  (60)

using the definitions of the mutual impedances

Z ₁₃ =jωk ₁₃√{square root over (L ₁ L ₃)},  (61)

Z ₂₃ =jω _(s) k ₂₃√{square root over (L ₂ L ₃)},  (62)

The changes in mutual impedance may be written in analogy e.g., toEquation (47)

ΔZ ₁₃=α₁₄α₃₄ Z ₄*,  (63)

ΔZ ₂₃=α₂₄α₃₄ Z ₄*,  (64)

where Z₄* denotes the conjugate complex impedance of the model of theobject 110. The scalar transformation factors are obtained in analogy toEquations (48) and (49) as

$\begin{matrix}{{\alpha_{14} = \frac{\omega_{s}M_{14}}{Z_{4}}},} & (65) \\{{\alpha_{34} = \frac{\omega_{s}M_{34}}{Z_{4}}},} & (66) \\{{\alpha_{24} = \frac{\omega_{s}M_{24}}{Z_{4}}},} & (67) \\{\alpha_{34} = {\frac{\omega_{s}M_{34}}{Z_{4}}.}} & (68)\end{matrix}$

Contemplating Equation (60) and following, it may be appreciated thatthere exists a current pair I₁ and I₂ providing virtually zero outputvoltage V_(3,0)=V_(13,0)+V_(23,0)≅0 in absence of the object 110.Provided that the second current source 308 can be precisely adjusted tothe required value in terms of both amplitude and phase, perfect voltagecancellation may be achieved. Phase adjustment may be needed ifmaterials 310 unequally affect the coupling factors k ₁₃ and k ₂₃ interms of the loss angle so that arg{k ₁₃}≠arg{k ₂₃} results.

The voltage cancellation effect may generally not apply to the change involtage as produced by the object 110, thus there may be a net outputvoltage ΔV₃=ΔV₁₃+ΔV₂₃≠0 in presence of the object 110. However, as forthe flux balanced mutual impedance sensing technique of FIGS. 6A and 6C,there may exist an area (blind spot) for object positions where the netvoltage response is below a threshold or even virtually zero, which maybe formally expressed as

ΔV ₃ =ΔZ ₁₃ I ₁ +ΔZ ₂₃ I ₂≅0  (69)

For certain objects (e.g., a coin) and for square-shaped sense coils512, 514, and 516, the blind spot may be found along a diagonal line ofsense coil 514 corresponding with the symmetry line (not shown in FIG.7a ) of the sense coil arrangement 710. Except of the blind spot, thesense coil arrangement 710 of FIG. 7A may provide sensitivity above theentire area spanned by the three sense coils 512, 514, and 516 and evenoutside this area to some extent.

In theory, the circuit 700 may provide an infinite sensitivity withrespect to the fractional change ΔV₃/|V_(3,0)|. In practice however,sensitivity may be limited by electrical and mechanical instability andnoise.

Contemplating Equation (60) and following, it may be appreciated thatthe circuit 700 potentially allows for angle-true impedance measurementfor purposes as previously discussed in connection with FIG. 3D, if thephase angle of the currents I₁ and I₂ are similar (arg{I₁}≅arg{I₂}).This condition may be satisfied if materials 310 similarly affect thecoupling factors k ₁₃ and k ₂₃ in terms of loss angle so that arg{k₁₃}≅arg{k ₂₃} results. Assuming arg{I₁}≅arg{I₂}, either the ratio V₃/I₁or the ratio V₃/I₂ may be used for determining the angle arg{Z₃} of theobject's 110 impedance Z₃. Alternatively, the voltage V₃ may be relatedto the average of the currents I₁ and I₂.

In some implementations of an object detection circuit 100, sense coils512, 514, and 516 may correspond e.g., to sense elements 106 a, 106 b,106 c, respectively, of the sense element array 106 with reference toFIG. 2A. In some implementations, sense coil arrangement 710 aretemporarily (e.g., sequentially) configured for flux compensated mutualimpedance sensing by three neighboring sense elements (e.g., senseelements 106 a, 106 b, 106 c) using a multiplexer circuitry as furtherdiscussed with reference to FIG. 11. A first sense coil arrangement 710may comprise sense coil 106 a as the secondary sense coil 514, sensecoil 106 b as the first primary sense coil 512, and the sense coil 106 cas the second primary sense coils 516. A second sense coil arrangement710 may comprise sense coil 106 b as the secondary sense coil 514, sensecoil 106 c as the first primary sense coil 512, and the sense coil 106 das the second primary sense coils 516, etc. Sense coil triplestemporarily configured in this way may be overlapping. Overlapping sensecoil arrangements 710 may reduce or eliminate blind spots as previouslydiscussed in connection with FIG. 7C, if the blind spots of theplurality of the sense coil arrangements 710 are non-overlapping(disjoint). It may be also appreciated that in such implementation, thenumber of sense coil arrangements 710 that can be potentially configuredis larger than the number N of sense elements of the array 106. Anycombination of three sense elements of the plurality of sense elements106 a, 106 b, . . . , 106 n may be potentially used in a sense coilarrangement 710.

FIGS. 7D to 7H are cut views illustrating various exemplaryimplementations of planar sense coil arrangements 710 as they may beused for flux balanced mutual impedance sensing.

FIG. 7D illustrates an implementation of a sense coil arrangement 710where sense coil 512, 514, and 516 are coplanar and adjacent. This sensecoil arrangement 710 may apply to an implementation using a plurality ofsubstantially equal sense elements 106 a, 106 b, . . . , 106 n eachincluding a single planar sense coil arranged in an array 106 withreference to FIG. 2A. Triples of neighboring sense coils (e.g., sensecoil 106 a, 106 b, 106 c) are configured in accordance with the circuit700 of FIG. 7A and analogously to the description of FIG. 5D.

FIG. 7E illustrates another exemplary implementation in analogy to FIG.5E where sense coils 512, 514, and 516 are partially overlapping. Thissense coil arrangement 710 may apply to an implementation using aplurality of substantially equal sense elements 106 a, 106 b, . . . ,106 n, each including a single planar sense coil arranged in an array106 having a first and a second plane and where the sense coils in thefirst plane are offset relative to the sense coils in the second planeby half of the width of a sense coil. Triples of neighboring sense coils(e.g., sense coil 106 a, 106 b, 106 c) may be temporarily (e.g.,sequentially) configured analogously to the description of FIG. 5E.Sense coil triples that are temporarily (e.g., sequentially) configuredmay be overlapping. It may be also appreciated that in suchimplementation or operation, the number of triple sense coilarrangements 510 that can be potentially configured is larger than thenumber N of sense elements of the array 106.

FIG. 7F illustrates a further implementation of a sense coil arrangement710 in analogy to FIG. 5F where sense coils 512, 514, and 516 are fullyoverlapping (on top of each other). This sense coil arrangement 710 mayapply to an implementation using a plurality of sense elements 106 a,106 b, . . . , 106 n arranged in an array 106. Each of the pluralitysense elements 106 a, 106 b, . . . , 106 n includes a triple of planarsense coils stacked on top of each other. This sense coil arrangement710 may be also considered as a trifilar winding structure with a firstwinding disposed in a first plane and a second and third windingdisposed in a second and third plane, respectively.

FIG. 7G illustrates yet another implementation of a sense coilarrangement 710 in analogy to FIG. 5G where sense coil 512, 514, and 516are coplanar and arranged inside of each other. This sense coilarrangement 710 may apply to an implementation using a plurality ofsense elements 106 a, 106 b, . . . , 106 n arranged in an array 106.Each sense element includes a triple of coplanar sense coils arrangedinside each other.

FIG. 7H illustrates yet a further implementation in analogy to FIG. 5Hwhere sense coil 512, 514, and 516 are coplanar and interleaved. Thissense coil arrangement 710 may apply to an implementation using aplurality of sense elements 106 a, 106 b, . . . , 106 n arranged in anarray 106. Each sense element includes a triple of interleaved coplanarsense coils. This sense coil arrangement 710 may be also considered as atrifilar winding structure with a first, a second, and a third windingin the same plane.

While the implementations of FIGS. 7D and 7E may provide a substantialnet response ΔV₃ in presence of the object 110, the implementations ofFIGS. 7E to 7H may not. This may be particularly true for animplementation based on FIG. 7F with a relatively small vertical spacingbetween the three sense coils 512, 514, and 516. In such implementation,the object 110 affects the mutual impedances Z₁₃ and Z₂₃ almost equally(ΔZ₁₃≅ΔZ₂₃). Since Z₁₃≅Z₂₃ and thus I₁≅−I₂, a very low net response ΔV₃may result in presence of the object 110 as evident from Equation (60).This may be true for any position of the object 110.

In some implementations of the object detection circuit 100 of FIG. 1based on the flux balanced mutual impedance sensing techniqueillustrated in FIG. 7A, the object detection circuit 100 is configuredto calibrate the mutual impedance measurement for purposes as previouslydescribed in connection with FIGS. 3A to 3D. However, implementation ofsuch calibration may require higher circuit complexity and cost incertain aspects, e.g., compared to aspects of capacitively compensatedself-impedance sensing using sense frequency tuning as discussed inconnection with FIG. 4A.

A technique that combines several advantages as previously discussedwith reference to FIGS. 4A and 5A and that avoids certain issues relatedto blind spots as previously discussed with reference to FIGS. 6A and 6Cis illustrated by the circuit 800 of FIG. 8A. This technique is referredto as capacitively compensated mutual impedance sensing. The circuit 800is also based on the current source voltage measurement approach. Asense coil arrangement 510 is used for sensing the object 110. The sensecoil arrangement 510 includes a primary sense coil 512 with inductanceL₁ and a secondary sense coil 514 with inductance L₂, each having afirst and second terminal. The circuit 800 further includes a capacitor820 (compensation or tuning capacitor) with capacitance C. The capacitor820 has a first and second terminal. The first terminal of the capacitor820 is electrically connected to the second terminals of each of thesense coils 512 and 514. Further, the circuit 800 includes analternating current source 306 that is tunable in frequency and thatdrives a current I₁ at sense frequency f_(s) into the primary sense coil512. The current source 306 is electrically connected to the firstterminal of the primary sense coil 512 and to the second terminal of thecapacitor 820 (in some implementations the second terminal of thecapacitor 820 is electrically connected to a reference groundpotential). The circuit 800 also includes a voltage measurement circuit304 configured to measure an output voltage V₂=V_(2,0)+ΔV₂. The voltagemeasurement circuit 304 is electrically connected to the first terminalof the secondary sense coil 514 and to the second terminal of thecapacitor 820. The voltage measurement circuit 304 may be frequencyselective (narrowband) tuned to the sense frequency f_(s). FIG. 8A alsoindicates presence of materials 310 by the shaded area.

An equivalent circuit of the circuit 800 of FIG. 8A including a circuitmodel of the object 110 is shown in FIG. 8B. In this equivalent circuit,the primary sense coil 512 and secondary sense coil 514 are eachrepresented by an equivalent inductance L₁ and L₂, respectively, and byan equivalent loss resistance R₁ and R₂. Though not shown in FIG. 8B,each of these elements may be a function of temperature ϑ. Theequivalent loss resistances R₁ and R₂ may each include a firstresistance component due to losses in the sense coil's conductivestructure (e.g., copper wires or PCB traces) and a second resistancecomponent due to loss effects in materials 310 as previously discussedin connection with FIG. 3B. Magnetic coupling between sense coils 512and 514 is represented by a complex coupling factor k ₁₂, which may bealso a function of temperature ϑ as previously explained in connectionwith FIGS. 5B and 5C. As also previously discussed, the equivalentinductances and resistance L₁, L₂, R₁ and R₂, respectively, as well asthe coupling factor k ₁₂ may include a change due to the presence ofmaterials 310. Since the electromagnetic properties of materials 310 maygenerally be temperature dependent, some temperature dependence may beexpected for the coupling factor k ₁₂. The change in k ₁₂ due tomaterials 310 may include a real and imaginary component due to reactiveand resistive (loss) effects, respectively, in materials 310 but alsodue to mutual loss effects in the sense coils 512 and 514 as previouslydiscussed. The real component relates to the mutual inductance M₁₂defined by Equation (29) while the imaginary component relates to anequivalent mutual resistance R₁₂ defined by Equation (31). FIG. 8B showseach of the primary sense coil 512 and the secondary sense coil 514magnetically coupled with coupling factor k₁₃ and k₂₃, respectively, tothe object 110 represented by an LR-Model (L₃, R₃) as previouslydiscussed e.g., with reference to FIG. 3B. The compensation capacitorappears with its equivalent capacitance C that may be also a function oftemperature ϑ depending on the type of capacitor as previously discussedin connection with FIG. 4A.

FIG. 8C shows a simplified equivalent circuit of the circuit 800 of FIG.8A. The sense coil arrangement 512 is represented by a simplified ‘T’equivalent circuit model neglecting first and second series branchimpedance as previously discussed with reference to FIG. 6B. Thissimplified equivalent circuit solely comprises the shunt branchimpedance that is now referred to as the capacitively compensated mutualimpedance Z_(12c). It includes the mutual inductance M₁₂, the equivalentmutual resistance R₁₂, the capacitance C that appears now in series tomutual inductance M₁₂, and the equivalent mutual impedance change ΔZ₁₂that represents the object 110 abstracted away.

Transimpedance as measured with the current source voltage measurementapproach is ideally only determined by the compensated mutual impedanceZ_(12,c) and may be expressed as

$\begin{matrix}{Z_{12\; c} = {{Z_{{12\; c},0} + {\Delta \; Z_{12}}} = {{R_{12} + {j\; \omega_{s}M_{12}} + \frac{1}{j\; \omega_{s}C} + {\Delta \; Z_{12}}} = {\frac{V_{2,0} + {\Delta \; V_{2}}}{I_{1}}.}}}} & (70)\end{matrix}$

Based on equation (70), it may be appreciated that there exists anangular sense frequency ω_(s) where the capacitor 820 perfectlycompensates for the mutual inductance M₁₂ (or the mutual reactanceω_(s)M₁₂) so that

Z _(12c,0) =R ₁₂  (71)

results. This condition may also be considered as some sort of resonanceand may be achieved by tuning the frequency f_(s) of the current sourceso that the magnitude of the mutual impedance Z_(12c) in absence of theobject 110 denoted by |Z_(12c,0)| becomes a minimum.

Assuming perfect compensation and using Equations (46) (47) and (52),the detection sensitivity of capacitively compensated mutual impedancesensing as defined by the fractional mutual impedance change may beexpressed as

$\begin{matrix}{\frac{\Delta \; Z_{12}}{Z_{{12\; c},0}} = {\frac{\alpha_{13}\alpha_{23}Z_{3}^{*}}{Z_{{12\; c},0}}.}} & (72)\end{matrix}$

Defining a Q-factor for the mutual impedance of the sense coilarrangement 510

$\begin{matrix}{{Q_{12} = \frac{\omega_{s}M_{12}}{R_{12}}},} & (73)\end{matrix}$

Equation (72) may be expressed in terms of Q₁₂, coupling factors k₁₃,k₂₃, |k ₁₂| and the object's 110 Q-factor Q₃ analogously to Equation(56) as follows:

$\begin{matrix}{\frac{\Delta \; Z_{12}}{Z_{{12\; c},0}} \cong {\frac{k_{13}k_{23}}{{\underset{\_}{k}}_{12}}\frac{Q_{12}Q_{3}}{1 + Q_{3}^{2}}{( {1 - {j\; Q_{3}}} ).}}} & (74)\end{matrix}$

For non-ferromagnetic objects 110 with high enough Q-factor Q₃>>1,equation (74) may be rewritten as

$\begin{matrix}{\frac{\Delta \; Z_{12}}{Z_{{12\; c},0}} \cong {\frac{k_{13}k_{23}}{{\underset{\_}{k}}_{12}}{Q_{12}( {\frac{1}{Q_{3}} - j} )}}} & (75)\end{matrix}$

showing that the fractional change |ΔZ₁₂|/|Z_(12c,0) that may beachievable with the circuit 800 of FIG. 8A using mutual reactancecompensation is Q₁₂-times higher than the fractional change|ΔZ₁₂|/|Z_(12,0)| obtained with the circuit 500 of FIG. 5A. Therefore,in some aspects of an object detection circuit 100, mutual reactancecompensation as illustrated by circuit 800 of FIG. 8A may be consideredan improvement over the inductive sensing technique as illustrated bycircuit 500 of FIG. 5A.

For the special case of a tightly coupled sense coil arrangement 510with |k ₁₂|≅<1 (e.g., using two identical (or substantially similar)sense coils 512 and 514 on top of each other and with zero displacementas shown by FIG. 5F, it may be a consequence that both sense coils 512and 514 provide about equal coupling to the object 110 (k₁₃≅k₂₃). Inthis special case, equation (75) may be rewritten as

$\begin{matrix}{\frac{\Delta \; Z_{12}}{Z_{{12\; c},0}} \cong {k_{13}^{2}{Q_{12}( {\frac{1}{Q_{3}} - j} )}}} & (76)\end{matrix}$

Further, using Equation (29) and the assumptions of |k ₁₂|≅<1 and twoidentical sense coils 512 and 514 (L₁=L₂), it follows that

M ₁₂ ≅L ₁ =L ₂  (77)

The Q-factor of the mutual impedance (Equation (73)) may then be alsoexpressed as

$\begin{matrix}{Q_{12} \cong {\frac{\omega_{s}L_{1}}{R_{12}}.}} & (78)\end{matrix}$

Moreover, assuming each of the sense coils 512 and 514 of the circuit800 of FIG. 8A identical (or substantially similar) to the sense coil302 of the circuit 400 of FIG. 4A, the same coupling k₁₃ to the object110, the same object 110, and same material 310 for both circuits 800and 400, a higher Q-factor Q₁₂ may be expected from the circuit 800relative to the Q-factor Q₁ of the circuit 400. Note that R₁₂ does notinclude the resistances of the sense coils' 512 and 514 windings so thatR₁₂<R₁ may be expected. However, apart from a resistance contributiondue to losses in materials 310, which may be inherent likewise to R₁₂and R₁, the equivalent mutual resistance R₁₂ may additionally include aresistance component from mutual loss effects e.g., an eddy current losseffect in the secondary sense coil 514 produced by the magnetic field ofthe primary sense coil 512. This loss resistance component may bediminished by keeping a small vertical distance between the two sensecoils 512 and 514 (e.g., in sense coil arrangement 510 of FIG. 5F) suchthat |k ₁₂|≅<1 remains valid. As such, in an aspect, a small verticaldistance between the two sense coils 512 and 514 is provided to diminishthe loss resistance. In reality, there may exist an optimum spacing forthe sense coils 512 and 514 maximizing Q₁₂. A gain in Q-factor by afactor of four was found in a non-optimized experimental set-upintegrated into a wireless power transfer structure 236 using identicalsense coils with equal length lead lines for all sense coils 512, 514,and 302.

Based on Equation (22) and (23) relating the fractional change of animpedance to temperature sensitivity, a potentially lower temperaturesensitivity may be expected from the capacitively compensated mutualimpedance sensing if compared to compensated impedance sensing (e.g.,circuit 400 of FIG. 4A).

In some implementations of the object detection circuit 100 of FIG. 1based on capacitively compensated mutual impedance sensing, mutualimpedance measurement may be subjected to measurement errors.Analogously to reactance compensation e.g., in the circuit 400 of FIG.A, mutual reactance compensation may provide a mechanism for accuratecalibration of the mutual impedance measurement e.g., with respect tothe angle arg{Z_(12c)} for purposes as previously discussed inconnection with the circuit 300 of FIG. 3A. In some exemplaryimplementations of an object detection circuit 100, the frequency f_(s)of the current source 306 is tuned such that the magnitude impedance|Z_(12c,0)| becomes substantially a minimum in absence of the object110, meaning that |Z_(12c,0)|≅R₁₂. Knowing that in absence of the object110, the minimum |Z_(12c,0)| ideally corresponds to a zero angle(arg{Z_(12c,0) }=0), the object detection circuit 100 may correct aspart of a calibration procedure the actually measured mutual impedanceby rotating the impedance plane so that the Im{Z_(12c,0)} vanishes.

As with compensation capacitor 420 in the circuit 400 of FIG. 4A,compensation capacitor 820 together with current source 306 and avoltage measurement circuit 304 each presenting a high input impedanceat sense frequency f_(s) and a low impedance at low frequencies may forma high pass filter to attenuate low frequency signal components e.g., atthe wireless power transfer frequency f_(wpt) as described in moredetail in connection with FIG. 12. This may result in relaxedrequirements for the voltage measurement circuit 304 and the currentsource 306 with respect to the dynamic range, overvoltage capability,etc. as previously discussed in connection with the circuit 400 of FIG.4A.

Multiplexed Implementations of Mutual Impedance Sensing

FIG. 9 is a block diagram of a circuit 900 illustrating exampleimplementations or operations of an object detection circuit 100. Theblock diagram may apply to any of the impedance and transimpedance(e.g., mutual impedance) sensing techniques and variants thereof asdescribed with reference to FIGS. 3A to 8A as well as to variousimpedance or transimpedance measurement approaches (e.g., the currentsource voltage measurement approach as previously described withreference to FIGS. 3A and 5A). The circuit 900 includes a driver circuit910, a control circuit 950 associated with the driver circuit 910, aplurality of sense circuits 104, a measurement circuit 920, and acontrol, processing & evaluation circuit 960 associated with themeasurement circuit 920.

The plurality of sense circuits 104 illustrated by the sense circuits104 a, 104 b, some dots and sense circuit 104 n may include any number N(e.g., N=64) of sense circuits. The plurality of sense circuits 104 isherein also referred to as the plurality of sense circuits 104 a, 104 b,. . . , 104 n. Each of the plurality of sense circuits 104 a, 104 b, . .. , 104 n includes a respective sense element of a plurality of senseelements 106 a, 106 b, . . . , 106 n as previously described withreference to FIG. 1. Each sense element may include one or more sensecoils (e.g., a sense coil arrangement 510 as it may be required by someof the mutual impedance sensing techniques as previously described withreference to FIGS. 5A to 8A). The plurality of sense elements 106 a, 106b, . . . , 106 n may be arranged in an array (e.g., array 106 aspreviously described with reference to FIGS. 1 and 2A). In someimplementations of the object detection circuit 100 of FIG. 1 usingcapacitive compensation of an impedance or a mutual impedance asdescribed with reference to FIGS. 4A and 8A, each of the plurality ofthe sense circuits 104 a, 104 b, . . . , 104 n may include acompensation (tuning) capacitor (e.g., capacitor 420 or 820 as shown inFIG. 4A or 8A, respectively). Each of the plurality of sense circuits104 a, 104 b, . . . , 104 n may be tuned substantially to a nominalsense frequency. In some implementations, there is a nominal sensefrequency common to each of the plurality of sense circuits 104 a, 104b, . . . , 104 n. In other implementations, sense circuits of theplurality of sense circuits 104 a, 104 b, . . . , 104 n areintentionally or unintentionally tuned to different nominal sensefrequencies.

The driver circuit 910 (e.g., a portion of the current source 306 withreference to FIG. 3A) is electrically connected to the plurality ofsense circuits 104 a, 104 b, . . . , 104 n. In some implementations, thedriver circuit 910 is configured to operate as a current source (e.g.,current source 306 as described in connection with FIG. 3A) andselectively (e.g., sequentially) apply a sense current signal I₁ (asindicated in FIG. 9) at an operating (sense) frequency f_(s) to each ofthe plurality of sense circuits 104 a, 104 b, . . . , 104 n. In otherimplementations, the driver circuit 910 is configured to operate as avoltage source e.g., as described in connection with FIG. 3A andselectively (e.g., sequentially) apply a sense voltage signal V₁ (asindicated in FIG. 9) at an operating (sense) frequency f_(s) to each ofthe plurality of sense circuits 104 a, 104 b, . . . , 104 n. In yetother implementations, the driver circuit 910 is configured to operateas a source that may be characterized neither by a current source nor avoltage source. For purposes of measuring an impedance or a mutualimpedance, the driver circuit 910 may generate a sinusoidal sense signalwith a defined frequency, amplitude and phase. The followingdescriptions of the circuit 900 of FIG. 9 assume a sinusoidal sensesignal. However, it should be appreciated that other sense signalwaveforms may be used in certain implementations as previously discussedin connection with FIG. 3A.

The driver circuit 910 may be configured to generate and apply more thanone sense signal (e.g., sense current signal I₁) at a time as it may berequired in some implementations or operations of an object detectioncircuit 100. In some implementations or operations, the concurrentlyapplied sense signals may generally differ in frequency and amplitude. Adriver circuit 910 capable of concurrently generating and applying aplurality of sense signals with different (distinct) sense frequenciesf_(s) may be used in some implementations or operations of the objectdetection circuit 100 e.g., to expedite an impedance measurement in eachof the plurality of sense circuits 104 a, 104 b, . . . , 104 n and/or toexpedite a transimpedance measurement in each of a plurality of sensecircuit pairs e.g., in an implementation where each sense circuitincludes a single sense coil (e.g., sense coil 512 or 514) as discussedwith reference to FIG. 5D.

A driver circuit 910 configured to provide two outputs (e.g., currentsource outputs) each delivering a sinusoidal signal (e.g., sense currentsignal I₁) with the same frequency f_(s) but generally differentamplitude and phase may be used to accomplish flux balanced mutualimpedance sensing as described with reference to FIG. 7A. An exampleimplementation of a driver circuit 910 configured to provide two currentsource outputs delivering respective current signals I_(1a) and I_(1b)is illustrated by FIG. 11.

A control circuit 950 electrically connected to the driver circuit 910is shown for illustrative purposes to indicate control of the drivercircuit 910.

The measurement circuit 920 is electrically connected to each of theplurality of sense circuits 104 a, 104 b, . . . , 104 n. In someimplementations, the measurement circuit 920 is configured toselectively (e.g., sequentially) measure an electrical voltage V₂ (asindicated in FIG. 9) in each of the plurality of sense circuits 104 a,104 b, . . . , 104 n and to provide a measurement output V_(out)proportional to the electrical voltage V₂. In some implementations, themeasurement output V_(out) is proportional to an electrical voltage V₂in one of the plurality of sense circuits 104 a, 104 b, . . . , 104 n inresponse to the at least one sense current signal I₁ being applied bythe driver circuit 910 to the respective at least one sense circuit ofthe plurality of sense circuits 104 a, 104 b, . . . , 104 n. In otherimplementations, the measurement circuit 920 is configured toselectively (e.g., sequentially) measure an electrical current I₂ (asindicated in FIG. 9) in each of the plurality of sense circuits 104 a,104 b, . . . , 104 n and to provide a measurement output V_(out)proportional to the electrical current I₂. In some implementations, themeasurement output V_(out) is proportional to an electrical current I₂in one of the plurality of sense circuits 104 a, 104 b, . . . , 104 n inresponse to the at least one sense voltage signal V₁ being applied bythe driver circuit 910 to the respective at least one sense circuit ofthe plurality of sense circuits 104 a, 104 b, . . . , 104 n.

In some implementations of an object detection circuit 100, themeasurement circuit 920 is an analog front-end portion of a more complexmeasurement circuit that also encompasses functions included in thecontrol, processing & evaluation circuit 960. In certainimplementations, the measurement circuit 920 may include a multiplexer(e.g., multiplexer 922) and an amplifier (e.g., amplifier 924) asillustrated in FIG. 9 by dashed lines.

In certain implementations or operations, the measurement circuit 920 isconfigured to measure more than one voltage V₂ or current I₂ at a timeand to provide more than one respective measurement output V_(out).Concurrent multiple voltage or current measurement may be required insome implementations or operations of the object detection circuit 100e.g., to expedite an impedance and/or a transimpedance and/or a fluxcompensated mutual impedance measurement in each of the plurality ofsense circuits 104 a, 104 b, . . . , 104 n and/or in pairs and/or evenin triples of sense circuits, respectively, of the plurality of sensecircuits 104 a, 104 b, . . . , 104 n as previously discussed in thecontext of the driver circuit 910. In certain implementations oroperations of an object detection circuit 100, measurement of aplurality of voltages V₂ or currents I₂ is performed concurrently andfrequency selectively at the respective frequency f_(s) of each of theplurality of sense signals I₁ or V₁ that is selectively applied to theplurality of sense circuits 104 a, 104 b, . . . , 104 n.

A control, processing & evaluation circuit 960 electrically connected tothe measurement circuit 920 is shown for illustrative purposes toindicate control of the measurement circuit 920 and the furtherprocessing.

The following descriptions on FIG. 9 assume example implementationsbased on the current source voltage measurement approach as describedwith reference to FIG. 3A. However, this should not exclude otherimpedance or transimpedance measurement approaches (e.g., voltage sourcecurrent measurement approach) that may apply.

As described above, the driver circuit 910 is configured to operate asat least one current source (e.g., characterized by a quasi-idealcurrent source as previously defined in connection with FIG. 3A) andselectively (e.g., sequentially) apply at least one sense current signalI₁ with a defined amplitude and phase at an operating frequency (sensefrequency) f_(s) to each of the plurality of sense circuits 104 a, 104b, . . . , 104 n. In one operational or implementational example, onlyone sense circuit of the plurality of sense circuits 104 a, 104 b, . . ., 104 n is driven at a time. To accomplish selective driving of each ofthe plurality of sense circuits 104 a, 104 b, . . . , 104 n, the drivercircuit 910 may include components such as at least one signal source912 (e.g., generating the sinusoidal sense signal with a definedfrequency f_(s), amplitude and phase), at least one driver amplifiercircuit 914, and at least one multiplexer circuit 916 also referred toas the input multiplexer. The at least one driver amplifier circuit 914is configured to provide a current source output characterized e.g., asa quasi-ideal current source suitable for selectively driving each ofthe plurality of sense circuits 104 a, 104 b, . . . , 104 n based on asignal received from the respective signal source 912. The at least onemultiplexer circuit 916 may be electrically connected between the outputof the respective driver amplifier circuit 914 and the plurality ofsense circuits 104 a, 104 b, . . . , 104 n and is configured toselectively connect each of the plurality of sense circuits 104 a, 104b, . . . , 104 n to the output of the respective driver amplifiercircuit 914. The driver circuit 910 may also return at least onereference signal to the control circuit 950. This at least one referencesignal may be representative for the respective sense current signal I₁in terms of frequency, amplitude, and phase. This at least one referencesignal may be used in the control, processing & evaluation circuit 960e.g., for computing at least one ratio V_(out)/I₁ (complex value) thatmay relate to an impedance or a transimpedance (e.g., a mutualimpedance).

The control circuit 950 is configured to apply one or more controlsignals to the driver circuit (e.g., to control the at least onemultiplexer circuit 916) to cause selective connection of the at leastone driver amplifier circuit 914 to each of the plurality of sensecircuits 104 a, 104 b, . . . , 104 n and to control parameters (e.g.,frequency, amplitude, and phase) of the at least one signal source 912.In certain implementations, the control circuit 950 is also configuredto receive the at least one reference signal representative for therespective sense current signal I₁ from the driver circuit 910. FIG. 9shows the control circuit 950 electrically linked with the control,processing & evaluation circuit 960. This electrical link will berequired in some implementations to exchange information between the twocircuits (e.g., to pass the at least one reference signal to thecontrol, processing & evaluation circuit 960).

As described above, the measurement circuit 920 is configured toselectively (e.g., sequentially) measure at least one electrical voltageV₂ in each of the plurality of sense circuits 104 a, 104 b, . . . , 104n and to provide at least one measurement output V_(out) proportional tothe respective electrical voltage V₂. In one operational orimplementational example, only one electrical voltage V₂ is measured ata time. The at least one input of the measurement circuit 920 may becharacterized by a quasi-ideal measurement circuit as defined withreference to FIG. 3A. To accomplish selective voltage measurement ineach of the plurality of sense circuits 104 a, 104 b, . . . , 104 n, themeasurement circuit 920 may include components such as at least onemultiplexer circuit 922 also referred to as the output multiplexer andat least one measurement amplifier circuit 924. The at least onemeasurement amplifier circuit 924 may be configured to provide asufficiently high input impedance as needed for a quasi-ideal voltagemeasurement e.g., as defined with reference to FIG. 3A and low noise inits measurement output V_(out). The at least one multiplexer circuit 922may be electrically connected between the input of the respectivemeasurement amplifier circuit 924 and the plurality of sense circuits104 a, 104 b, . . . , 104 n and is configured to selectively connecteach of the plurality of sense circuits 104 a, 104 b, . . . , 104 n tothe input of the respective measurement amplifier circuit 922.

The control, processing & evaluation circuit 960 is configured to applyone or more control signals to the measurement circuit 920 (e.g., tocontrol the at least one multiplexer circuit 922) to cause selectiveconnection of the at least one measurement amplifier circuit 924 to eachof the plurality of sense circuits 104 a, 104 b, . . . , 104 n. Further,it is configured to provide further signal processing and evaluation ofthe acquired measurement data. The control, processing & evaluationcircuit 960 may be configured to provide analog-to-digital signalconversion, frequency selective filtering, synchronous detection,combining outputs, summation of outputs, averaging of outputs, scalingof outputs, correction of outputs, evaluation of sequences (time-series)and/or patterns of outputs indicative of an impedance and/ortransimpedance (e.g., mutual impedance), etc. and eventually to decidewhether an object (e.g., object 110 of FIG. 3A) is proximate to at leastone sense element of the plurality of sense elements 106 a, 106 b, . . ., 106 n based on a change in an impedance or transimpedance (e.g.,mutual impedance) and to output at least a detection hypothesis H_(out)as indicated in FIG. 9.

The detection hypothesis H_(out) may be indicative for presence orabsence of an object (e.g., object 110) and may be used by the wirelesspower transfer system 200 with reference to FIG. 2A to control wirelesspower transfer. A positive detection hypothesis H_(out) may beindicative for presence of an object (e.g., object 110) and may causethe wireless power transfer system 200 to cease wireless power transferor to reduce a level of power. Conversely, a negative detectionhypothesis H_(out) may be indicative for absence of an object and maycause the wireless power transfer system 200 to continue or resumewireless power transfer or to return to the ordinary power level. Insome implementations, the detection hypothesis output H_(out) mayinclude additional information (e.g., the position of the at least onesense element associated to a positive hypothesis output H_(out) and/ora detection confidence value).

In some implementations or operations of a control, processing &evaluation circuit 960, the evaluation of measurement outputs is basedon the at least one measurement output V_(out) proportional to therespective electrical voltage V₂ in response to a current I₁. In otherimplementations or operations, the evaluation is based on the at leastone measurement output V_(out) in relation to the respective (known)current I₁ e.g., on the ratio V_(out)/I₁ (complex value) that may beindicative of an impedance or transimpedance (e.g., a mutual impedance).

In an implementation or operation of the object detection circuit 100 ofFIG. 1 employing an impedance sensing technique with reference to FIG.3A or 4A, the input multiplexer 916 and the output multiplexer 922 areconfigured and controlled in a manner to selectively (e.g.,sequentially) apply the current I₁ to a sense circuit (e.g., sensecircuit 104 a) of the plurality of sense circuits 104 a, 104 b, . . . ,104 n and to selectively (e.g., sequentially) measure the electricalvoltage V₂ in the same sense circuit (e.g., sense circuit 104 a). Inthis case, the ratio V_(out)/I₁ is indicative of the impedance of therespective sense circuit (e.g., sense circuit 104 a). The maximum numberN_(imp) of (self-) impedance measurements that may be performed equalsthe number N of the plurality of sense circuits 104 a, 104 b, . . . ,104 n. More formally,

N _(imp) =N.  (79)

In another implementation or operation of the object detection circuit100 of FIG. 1 using a sense element array (e.g., array 106 withreference to FIG. 1) and employing a transimpedance (e.g., mutualimpedance) sensing technique e.g., with reference to FIG. 5A, the inputmultiplexer 916 and the output multiplexer 922 are configured andcontrolled in a manner to selectively apply the current I₁ to a firstsense circuit (e.g., sense circuit 104 a) of the plurality of sensecircuits 104 a, 104 b, . . . , 104 n and to selectively measure theelectrical voltage V₂ in a second sense circuit (e.g., sense circuit 104b) different from the first sense circuit. In this case, the ratioV_(out)/I₁ is indicative of the transimpedance between the first andsecond sense circuit. This operation may apply e.g., in animplementation where each of the plurality of sense circuits 104 a, 104b, . . . , 104 n includes a sense element including a single sense coil(e.g., sense coil 302). Using combinatorics, the maximum numberN_(trans) of transimpedance measurements that may be performed in pairsof sense circuits of the plurality (N) of sense circuits 104 a, 104 b, .. . , 104 n may be formally expressed as

$\begin{matrix}{N_{trans} = {\begin{pmatrix}N \\2\end{pmatrix} = {\frac{N( {N - 1} )}{2}.}}} & (80)\end{matrix}$

This number includes transimpedance measurements between neighboring andnon-neighboring sense coils. In an example implementation of the objectdetection circuit 100 using a number N=64 sense circuits, each includinga single sense coil, the maximum number of transimpedance measurementsthat may be performed in pairs of sense circuits amounts according toEquation (83) to N_(trans)=2,016.

In a further implementation or operation of the object detection circuit100 of FIG. 1 employing a flux compensated mutual impedance sensingtechnique e.g., with reference to FIG. 7A, the input multiplexer 916 andoutput multiplexer 922 are configured and controlled in a manner toselectively apply a first current I₁ to a first sense circuit (e.g.,sense circuit 104 a) and a second current I₁ to a second sense circuit(e.g., sense circuit 104 b) and to selectively measure the electricalvoltage V₂ in a third sense circuit (e.g., sense circuit 104 c) of theplurality of sense circuits 104 a, 104 b, . . . , 104 n. In this case, achange in the measurement output V_(out) due to presence of an object(e.g., object 110) is indicative of a change in transimpedance betweenthe first sense circuit (e.g., sense circuit 104 a) and the third sensecircuit (e.g., sense circuits 104 c) and/or between the second sensecircuit (e.g., sense circuit 104 b) and the third sense circuit (e.g.,sense circuits 104 c). This operation may apply in an implementationwhere each of the plurality of sense elements 106 a, 106 b, . . . , 106n consists of a single sense coil (e.g., sense coil 302). Usingcombinatorics, the maximum number N_(fcti) of flux compensated mutualimpedance measurements that may be performed in triples of sensecircuits of the plurality (N) of sense circuits 104 a, 104 b, . . . ,104 n may be formally expressed as

$\begin{matrix}{N_{fcti} = {\begin{pmatrix}N \\3\end{pmatrix} = {\frac{{N( {N - 1} )}( {N - 2} )}{3}.}}} & (81)\end{matrix}$

This number includes flux compensated mutual impedance measurementsbetween neighboring and non-neighboring sense coils. In an exampleimplementation of the object detection circuit 100 using a number N=64of sense circuits, each including a single sense coil, the maximumnumber of flux compensated transimpedance measurements that may beperformed in triples of sense circuits amounts according to Equation(81) to N_(fcti)=83,328.

In some implementations or operations using a plurality of sensecircuits, each including a single sense coil, the object detectioncircuit 100 performs only a subset of all possible transimpedance orflux compensated mutual impedance measurements as described above. In anexample operation, transimpedance or flux compensated mutual impedancemeasurements may be limited to pairs or triples, respectively, ofneighboring sense coils.

In some aspect, the object detection circuit 100 performs an impedancemeasurement for each of the plurality of sense circuits 104 a, 104 b, .. . , 104 n and additionally, a transimpedance measurement in each of aplurality of pairs (or triples) of sense circuits of the plurality ofsense circuits 104 a, 104 b, . . . , 104 n to determine whether anobject (e.g., object 110) is in proximity of at least one of a sensecoil of the plurality of sense coils 106 a, 106 b, . . . , 106 n. It maybe appreciated that these additional transimpedance measurements mayprovide supplementary information useful to improve a detectionsensitivity or reliability as further discussed below.

In implementations or operations using a plurality of sense circuits 104a, 104 b, . . . , 104 n, each including a single sense coil, performingimpedance and transimpedance measurements in a time-multiplexed fashionmay be time consuming and may result in object detection latency. Thismay be even true if the object detection circuit 100 is configured toselectively and concurrently apply multiple sense current signals I₁ tothe plurality of sense circuits 104 a, 104 b, . . . , 104 n and toselectively and concurrently measure multiple electrical voltage V₂ inresponse to the current signals I₁ to expedite impedance andtransimpedance measurement as previously discussed. Therefore, incertain implementations or operations, the object detection circuit 100is configured to perform impedance measurements in times where objectdetection latency is critical (e.g., during active wireless powertransfer) and to perform additional transimpedance measurements in timeswhere object detection latency is uncritical (e.g., when wireless powertransfer is inactive).

In certain implementations of an object detection circuit 100, themeasurement output V_(out) as provided by the measurement circuit 920 isan analog signal e.g., a sinusoidal signal with an amplitudeproportional to the amplitude of the electrical voltage V₂. This analogmeasurement output V_(out) is further processed (e.g., digitized,filtered, evaluated, etc.) in the control, processing & evaluationcircuit 960 as previously discussed. Therefore, in some cases,measurement outputs V_(out) may also refer to outputs as produced in thecontrol, processing & evaluation circuit 960. These outputs are notshown in FIG. 9.

In an example implementation or operation of a control, processing &evaluation circuit 960, the evaluation is based on an absolute detectionscheme, where at least one measurement outputs V_(out) of a plurality ofmeasurement outputs, each associated to at least one sense circuit(e.g., sense circuit 104 a or pair of sense circuits 104 a and 104 b) ofthe plurality of sense circuits 104 a, 104 b, . . . , 104 n is comparedagainst a respective reference value V_(out,0). The at least onemeasurement output V_(out) may be indicative of an impedance and/or atransimpedance (e.g., a mutual impedance) and may refer to an output asobtained after processing (e.g., filtering, combining, averaging, etc.)in the control, processing & evaluation circuit 960. The respectivereference value V_(out,0) may be the measurement output V_(out)associated to the same at least one sense circuit (e.g., sense circuit104 a or pair of sense circuits 104 a and 104 b) in absence of theobject 110 and may have been determined in a process of calibration. Insome implementations or operations based on an absolute detectionscheme, presence of an object (e.g., object 110) is assumed, if at leastone difference between a measurement output V_(out) associated to the atleast one sense circuit (e.g., sense circuit 104 a or pair of sensecircuits 104 a and 104 b) and the respective reference value V_(out,0)exceeds a threshold. This difference may be indicative of the change ΔZ₁in an impedance Z₁ or to a change ΔZ₁₂ in a mutual impedance Z₁₂ aspreviously defined e.g., with reference to FIGS. 3A and 5A,respectively.

In an example implementation or operation of a control, processing &evaluation circuit 960, the evaluation is based on a time-differentialdetection scheme that is sensitive e.g., to a fast (e.g., abrupt) changein a sequence (time-series) of consecutive measurement outputs V_(out),each associated to the same at least one sense circuit (e.g., sensecircuit 104 a or pair of sense circuits 104 a and 104 b) of theplurality of sense circuits 104 a, 104 b, . . . , 104 n. The measurementoutputs V_(out) may be indicative of an impedance and/or transimpedance(e.g., a mutual impedance) and may refer to outputs as obtained afterprocessing (e.g., filtering, combining, averaging, etc.) in the control,processing & evaluation circuit 960. In some implementations oroperations based on a time-differential detection scheme, presence of anobject (e.g., object 110) is assumed, if at least one difference betweena measurement output V_(out) associated to at least one sense circuit(e.g., sense circuit 104 a or pair of sense circuits 104 a and 104 b) ofthe plurality of sense circuits 104 a, 104 b, . . . , 104 n and to atleast one first time and a measurement output V_(out) associated to thesame at least one sense circuit and at least one second time exceeds athreshold. Using time-differential detection, an object can potentiallybe detected when it enters or leaves the proximity of the at least onesense element (e.g., sense element 106 a) or generally when it moves inthe proximity of the at least one sense element.

In a further example implementation or operation of a control,processing & evaluation circuit 960, the evaluation is based on a sensecircuit-differential detection scheme that is sensitive to differencesbetween measurement outputs V_(out) associated to different sensecircuits or different pairs of sense circuits of the plurality of sensecircuits 104 a, 104 b, . . . , 104 n. This detection scheme may be alsoreferred to as space-differential detection. The measurement outputsV_(out) may be indicative of an impedance and/or a transimpedance (e.g.,a mutual impedance) and may refer to outputs as obtained afterprocessing (e.g., filtering, combining, averaging, etc.) in the control,processing & evaluation circuit 960. In some implementations oroperations based on a space-differential detection scheme, presence ofan object (e.g., object 110) is assumed if at least one differencebetween a measurement output V_(out) associated to at least one firstsense circuit (e.g., sense circuit 104 a or pair of sense circuits 104 aand 104 b) of a plurality of sense circuits 104 a, 104 b, . . . , 104 nand a measurement output V_(out) associated to at least one second sensecircuit (e.g., sense circuit 104 b) exceeds a threshold. In someimplementations or operations of a true space-differential detectionscheme, the plurality of measurement outputs V_(out) used to determineat least one difference refer to substantially the same time. It may beappreciated that in certain cases, space-differential detection may beless sensitive and reliable than time-differential detection since sensecircuits of the plurality of sense circuits 104 a, 104 b, . . . , 104 nmay be at least partially differently (individually) affected bytemperature, mechanical impacts, and ageing.

For certain implementations of the object detection circuit 100 of FIG.1 solely relying on an absolute detection scheme, calibration forpurposes of determining reference values as previously mentioned may beimportant. Moreover, absolute detection may require a circuit 900 eitherwith high electrical and mechanical long-term stability with respect totemperature variations, mechanical impacts, and ageing or alternatively,high sensitivity (e.g., high fractional impedance change as previouslydefined with reference to FIG. 3A). For implementations of the objectdetection circuit 100 employing a time-differential or aspace-differential detection approach or a combination thereof,calibration and long-term stability may be less important.

In some aspects, time-differential detection may be sensitive tomovements of metallic structures in the environment of the sense elementarray (e.g., array 106). Such environmental effects may includemicromovements of materials (e.g., materials 310 with reference to FIG.3A) inside the wireless power transfer structure 224 that integrates thesense element array. Further, environmental effects may includemovements of the metallic vehicle underbody structure including thevehicle-side wireless power transfer structure 260 when a vehicle isparked over the wireless power transfer structure 224. Vehicle underbodymovements may be caused e.g., by the vehicle suspension system thatbounces when persons are entering or leaving the vehicle or due to windforces acting on the vehicle body. Disturbance effects from movements ofmetallic structure in the environment such as the vehicle underbody maycause false detections in certain implementations of the objectdetection circuit 100 that is solely based on a time-differentialapproach. Therefore, in some aspects, it may be desirable to mitigatesuch disturbance effects. Combining the time-differential scheme with aspace-differential scheme on top may be an approach to effectivelydiscriminate such disturbance effects. In a space-differential approach,presence of an object (e.g., object 110) is determined based ondifferences between measurement outputs V_(out) associated to differentsense circuits or pairs of sense circuits of the plurality of sensecircuits 104 a, 104 b, . . . , 104 n as previously discussed. In certainimplementations or operations of a space-differential detection scheme,presence of an object is determined by evaluating at least onedifference between a measurement output V_(out) associated to at leastone sense circuit (e.g., sense circuit 104 a or pair of sense circuits104 a and 104 b) and a reference value that is determined based on aplurality of measurement outputs V_(out), each associated to a differentsense circuit or a different pair of sense circuits of the plurality ofsense circuits 104 a, 104 b, . . . , 104 n. This reference value may bee.g., an arithmetic mean value, a r.m.s. value, a median value (50^(th)percentile), or any other percentile value that is derived from ahistogram built upon the plurality of measurement outputs V_(out). Itmay be appreciated that this special space-differential scheme has thepotential to discriminate environmental effects e.g., from a movingvehicle underbody that may produce time-varying impedance or mutualimpedance changes in a majority (cluster) of sense circuits or pairs ofsense circuits, respectively. This special scheme may be considered as amechanism that automatically adapts the decision threshold used by thecontrol, processing & evaluation circuit 960 for determining presence ofan object (e.g., object 110). More specifically, in some implementationsor operations, the control, processing & evaluation circuit 960automatically adjusts the reference value as described above. When thevehicle underbody is moving, the reference value may rise and thus thedetection threshold. Increasing the detection threshold will reduce thefalse detection rate but also the detection sensitivity (detectionprobability) to some extent. Therefore, a somewhat lower sensitivitymust be accepted for an object entering the predetermined space whilethe vehicle is moving. As soon as the vehicle underbody comes to rest,the detection threshold is readjusted automatically, and the objectdetection circuit 100 may return to its ordinary sensitivity (detectionprobability) maintaining a specified false detection rate.

In yet a further example implementation or operation of a control,processing & evaluation circuit 960, the evaluation is based on apattern recognition approach (e.g., based on machine learningprinciples). In some implementations or operations, pattern recognitionmay be considered as a form of space-differential detection. A pluralityof measurement outputs V_(out) associated to different sense circuits ordifferent pairs of sense circuits or even different triples of sensecircuits (e.g., with reference to FIG. 7A) of the plurality of sensecircuits 104 a, 104 b, . . . , 104 n may be imaged as a pattern (e.g., a2D pattern or a 3D pattern). The measurement outputs V_(out) may beindicative of an impedance and/or a transimpedance (e.g., a mutualimpedance) and may refer to outputs V_(out) as obtained after processing(e.g., filtering, combining, averaging, etc.) in the control, processing& evaluation circuit 960. If the plurality of measurement outputsV_(out) include outputs indicative of a transimpedance (e.g., a mutualimpedance) obtained with a sense coil arrangement 510 or 710 aspreviously described with reference to FIG. 5D or 7D, respectively, thenumber N_(m) of measurement outputs V_(out) may be larger than thenumber N of the plurality of sense circuits 104 a, 104 b, . . . , 104 nas previously discussed with reference to Equations (79) and (80). Insome implementations or operations, the control, processing & evaluationcircuit 960 is trained with various objects placed at various positionsand in various orientations. It may be also trained for typicalenvironmental effects such as temperature variation, mechanical impacts,and ageing and for effects from the vehicle underbody structureincluding the vehicle-side wireless power transfer structure 260 withreference to FIG. 2B. An object (e.g., object 110) in proximity of atleast one sense element (e.g., sense element 106 a) of the plurality ofsense elements 106 a, 106 b, . . . , 106 n may produce a characteristicchange in the pattern of measurement outputs V_(out). In someimplementations or operations of a control, processing & evaluationcircuit 960 employing pattern recognition, presence of an object (e.g.,object 110) is assumed if a pattern resembles patterns learned in atraining with a variety of objects as previously described. In someimplementations of a pattern recognition approach, the control,processing & evaluation circuit 960 includes a neuronal networkconfigured and trained for detecting objects (e.g., object 110). Thisneuronal network may be also configured and trained to mitigate effectsfrom temperature variations and other environmental impacts aspreviously discussed.

Pattern recognition may be also contemplated as detecting a change of avector in a multi-dimensional vector space. The plurality (N_(m)) ofmeasurement outputs V_(out) each associated to a different sense circuitor to a different pair of sense circuits of the plurality of sensecircuits 104 a, 104 b, . . . , 104 n may be considered as a vector{right arrow over (V)}_(out) in a N_(m)-dimensional vector space. If anobject (e.g., object 110) is in proximity to at least one sense element(e.g., sense element 106 a), the vector {right arrow over (V)}_(out,0)as measured in absence of the object will generally experience a change{right arrow over (ΔV)}_(out). It may be found that changes {right arrowover (ΔV)}_(out) produced by objects (e.g., object 110) move in certain(typical) directions or in a certain subspace of the N_(m)-dimensionalvector space while changes produced by environmental effects aspreviously discussed move mostly in other directions or in anothersubspace. In a probabilistic sense, these two subspaces may be largelynon-overlapping (orthogonal). This orthogonality of the two subspacesmay improve when the number N_(m) increases. In some implementations ofan object detection circuit 100, the number N_(m) (dimensionality of thevector space) is increased e.g., by additionally measuring the mutualimpedance between multiple pairs of sense circuits of the plurality ofsense circuits 104 a, 104 b, . . . , 104 n. It may be appreciated thateach mutual impedance measurement that is added may provide additionalinformation that may potentially improve an object detection circuit100. Considering environmental effects such as from temperaturevariations, mechanical impacts, and ageing, it may be found thatincreasing the dimensionality N_(m) of the vector space is moreeffective in terms of detection reliability than increasing asensitivity (e.g., by increasing a fractional change, a SNR, or ameasurement time).

In some implementations of an object detection circuit 100, the control,processing & evaluation circuit 960 includes functions as needed forspecialized detection schemes. In certain implementations, it includese.g., a correlator function for purposes as discussed in more detailbelow.

Inductive thermal sensing may be a specialized detection scheme fordetecting a metallic object (e.g., object 110) by its temperaturevariation as occurring when exposed to a magnetic field (e.g., the lowfrequency magnetic field as generated by the wireless power transfersystem 200 with reference to FIG. 2A) with a level sufficient to producea substantial temperature variation. In some implementations oroperations of inductive thermal sensing, the change (response) in anelectrical characteristic (e.g., impedance Z₁, mutual impedance Z₁₂) andthus in the measurement output V_(out) due to the presence of themetallic object is stimulated by intermittently applying the magneticfield exposure in a manner so that the object's temperature is followingthe exposure ON and OFF cycles distinctly but not exceeding a criticallevel. This technique relies on an object's electrical properties (e.g.,conductivity, permeability) and thus its equivalent impedance Z₃ (withreference to FIG. 3B) being generally a function of the object'stemperature. It may be appreciated that the change (response) in anelectrical characteristic (e.g., impedance Z₁, mutual impedance Z₁₂) andthus in the measurement output V_(out) produced by an object whoseequivalent impedance Z₃ is a function of the object's temperature willgenerally also be a function of the object's temperature. In someimplementations or operations, presence of an object (e.g., object 110)is determined based on a correlation between the at least one firstsequence (time-series) of consecutive measurement outputs V_(out) eachassociated to the same at least one sense circuit (e.g., sense circuit104 a or pair of sense circuits 104 a and 104 b) of the plurality ofsense circuits 104 a, 104 b, . . . , 104 n and a corresponding secondsequence indicative of the level (e.g., the r.m.s. envelope) of thetime-varying magnetic field exposure signal.

Inductive thermal sensing may potentially provide a solution fordetecting objects of a certain category on an absolute basis as definedabove. This category may include metallic objects with a sufficienttemperature coefficient, strong eddy current heating effect when exposedto the magnetic field, and low thermal capacity so that they heat up andcool down faster than other metallic structures e.g., inside thewireless power transfer structure 224 or in its environment when exposedto the magnetic field. This category may include pieces of foil(Aluminum foil), metallized paper or the like. Non-limiting examples ofobjects belonging to this category may be a cigarette packet including ametallized foil, the cover of a yoghurt cup, a chewing gum wrapper, anda cigarette lighter with a metallized lighter head. Objects of thiscategory may heat up rapidly to temperatures e.g., above 500 K ifexposed to an alternating magnetic field with a flux density above 1 mTat a frequency of 85 kHz. Objects with temperatures above 500 K may beconsidered a potential risk for fire if the object comes into contactwith a flammable material such as paper, dry foliage, oil, fuel, etc.

In an example implementation or operation of the object detectioncircuit 100 of FIG. 1 employing inductive thermal sensing, the objectdetection circuit 100 controls the wireless power transfer system 200with reference to FIG. 2A to intermittently apply an exposure magneticfield (e.g., at a wireless power transfer frequency of 85 kHz) with alevel sufficient to produce a substantial temperature variation in ametallic object (e.g., object 110). While the exposure magnetic field isintermittently applied, the control, processing & evaluation circuit 960processes at least one first sequence of consecutive measurement outputsV_(out) each associated to the same at least one sense circuit (e.g.,sense circuit 104 a or pair of sense circuits 104 a and 104 b) todetermine a level of correlation between the at least one first sequenceof consecutive measurement outputs V_(out) and a corresponding secondsequence indicative of the level of the intermittently applied exposuremagnetic field signal. Presence of an object (e.g., object 110) isassumed if the level of correlation exceeds a threshold for at least onesense circuit or at least one pair of sense circuits of the plurality ofsense circuits 104 a, 104 b, . . . , 104 n.

In some example implementations or operations of an object detectioncircuit 100, inductive thermal sensing is combined with aspace-differential detection scheme. Presence of an object (e.g., object110) may be assumed if at least one difference between the correlatoroutput associated to at least one first sense circuit (e.g., sensecircuit 104 a) and the correlator output associated to at least onesecond sense circuit (e.g., sense circuit 104 b) exceeds a threshold.

Inductive ferromagnetic sensing may be another specialized detectionscheme for detecting a metallic ferromagnetic object (e.g., object 110)by the variation (modulation) of its electrical properties (e.g.,permeability, conductivity) as it may occur when exposed to a magneticfield (e.g., a low frequency magnetic field as generated by the wirelesspower transfer system 200 with reference to FIG. 2A) with a levelsufficient to produce a substantial variation. In some exampleimplementation or operation of inductive ferromagnetic sensing, thechange (response) in an electrical characteristic (e.g., impedance Z₁,mutual impedance Z₁₂) and thus in the measurement output V_(out) due tothe presence of the ferromagnetic object is stimulated by intermittentlyapplying the magnetic field exposure in a manner so that the object'selectrical properties are substantially modulated by the exposure ON andOFF cycles but also by the oscillation of the exposure magnetic field(e.g., with a frequency of 85 kHz) but with a level not causing theobject to exceed a critical temperature. This technique relies on anobject's electrical properties (e.g., conductivity, permeability) andthus its equivalent impedance Z₃ (with reference to FIG. 3B) beinggenerally a function of the instantaneous amplitude of the exposuremagnetic field signal. It may be appreciated that the change (response)in an electrical characteristic (e.g., impedance Z₁, mutual impedanceZ₁₂) and thus in the measurement output V_(out) produced by an objectwhose equivalent impedance Z₃ is a function of the instantaneousamplitude of the exposure magnetic field signal will generally also be afunction of the instantaneous amplitude of the exposure signal. Thisfunction may be a non-linear function, may include memory (hysteresis)effects, and may also include thermal effects (e.g., Curie temperatureeffect). In some implementations or operations, presence of an object isdetermined based on a correlation between at least one first sequence(time-series) of consecutive measurement outputs V_(out), eachassociated to the same sense circuit (e.g., sense circuit 104 a) of theplurality of sense circuits 104 a, 104 b, . . . , 104 n and acorresponding second sequence indicative of the exposure magnetic fieldsignal.

Inductive ferromagnetic sensing may potentially provide a solution fordetecting objects of a certain category on an absolute basis as definedabove. This category may include metallic ferromagnetic objects whoseelectrical properties substantially vary when exposed to a time-varying(alternating) magnetic field and whose response in a measured electricalcharacteristic is stronger than a response from any ferromagneticstructure inside the wireless power transfer structure 224 or in itsenvironment when exposed to the alternating magnetic field. Non-limitingexamples of objects belonging to this category may be paper clips,pieces of steel wire, steel nails, steel pins, screws, nuts, andwashers. Objects of this category may heat up to temperatures e.g.,above 500 K if exposed to an alternating magnetic field with a fluxdensity above 1 mT at a wireless power transfer frequency of 85 kHzsince ferromagnetism (permeability μ_(r)>1) increases the heating effectdue to the reduced skin depth (see Equation (15)) and additionalhysteresis loss effects.

In an example implementation or operation of the object detectioncircuit 100 of FIG. 1 using inductive ferromagnetic sensing, the objectdetection circuit 100 controls the power conversion circuit 222 withreference to FIG. 2A to intermittently apply an exposure magnetic fieldwith a level sufficient to produce a substantial variation in anelectrical property of the metallic ferromagnetic object (e.g., object110). While the exposure magnetic field is intermittently applied, thecontrol, processing & evaluation circuit 960 processes at least onesequence (time-series) of consecutive measurement outputs V_(out) eachassociated to the same at least one sense circuit (e.g., sense circuit104 a or pair of sense circuits 104 a and 104 b) of the plurality ofsense circuits 104 a, 104 b, . . . , 104 n to determine a level ofcorrelation between the at least one first sequence of consecutivemeasurement outputs V_(out) and a corresponding second sequenceindicative of the applied exposure magnetic field signal. Presence of anobject (e.g., object 110) is assumed if the level of correlation exceedsa threshold for at least one sense circuit or at least one pair of sensecircuits of the plurality of sense circuits 104 a, 104 b, . . . , 104 n.

In an example implementation or operation of an object detection circuit100, inductive ferromagnetic sensing is combined with aspace-differential detection scheme. Presence of an object (e.g., object110) is assumed if at least one difference between the correlator outputassociated to at least one first sense circuit (e.g., sense circuit 104a) and the correlator output associated to at least one second sensecircuit (e.g., sense circuit 104 b) exceeds a threshold. In anotherexample implementation or operation, respective correlator outputsobtained with inductive ferromagnetic sensing and with inductive thermalsensing are combined.

Inductive kinematic sensing may be a further specialized detectionscheme for detecting an object (e.g., object 110) by its relative motione.g., when the object is mechanically moved relative to the senseelements (e.g., the plurality of sense elements 106 a, 106 b, . . . ,106 n). In an example implementation or operation of inductive kinematicsensing, the top surface of the housing 236 of the wireless powertransfer structure 224 where an object may rest is mechanically movedback and forth relative to the sense element array 106. (Either thetop-surface of housing 236 is moved or the wireless power transferstructure including sense element array 106 inside housing 236 is moved,or both are moved.) In some implementations or operations of inductivekinematic sensing, an object is moved relative to the sense elementswith an amplitude e.g., in the centimeter range and at a low frequency(e.g., 3 Hz). In other implementations or operations, the object ismoved (vibrated) with an amplitude e.g., in the millimeter or evensubmillimeter range at a higher frequency (e.g., 50 Hz). The relativemovement of an object (e.g., object 110) in proximity of a sense element(e.g., sense element 106 a) may produce a time-varying change in anelectrical characteristic (e.g., impedance Z₁, mutual impedance Z₁₂) andthus in the measurement output V_(out). It may be appreciated that thischange (response) in an electrical characteristic will generally be afunction of the object's instantaneous position relative to the senseelement array 106. This function may be a non-linear function. In someimplementations or operations, presence of an object (e.g., object 110)is determined based on a correlation between the at least one firstsequence of consecutive measurement outputs V_(out) each associated tothe same at least one sense circuit (e.g., sense circuit 104 a or pairof sense circuits 104 a and 104 b) of the plurality of sense circuits104 a, 104 b, . . . , 104 n and a corresponding second sequenceindicative of the mechanical movement signal.

Inductive kinematic sensing may potentially provide a solution fordetecting objects of any category e.g., as described with reference toFIG. 3D on an absolute basis as defined above.

In an example implementation of a wireless power transfer system 200including the object detection circuit 100 of FIG. 1 employing inductivekinematic sensing, the wireless power transfer structure 224 includeselectrical and mechanical functions as required to move (or vibrate) apotential object (e.g., object 110 resting on the surface of housing236) relative to the sense element array 106. Moreover, the objectdetection circuit 100 may control the wireless power transfer structure224 with reference to FIG. 2B to apply mechanical movements with anamplitude and frequency as required for inductive kinematic sensing.While mechanical movements are applied, the control, processing &evaluation circuit 960 processes at least one first sequence ofconsecutive measurement outputs V_(out) each associated to the same atleast one sense circuit (e.g., sense circuit 104 a or pair of sensecircuits 104 a and 104 b) of the plurality of sense circuits 104 a, 104b, . . . , 104 n to determine a level of correlation between the atleast one first sequence of consecutive measurement outputs V_(out) anda corresponding second sequence indicative of the mechanical movementsignal. Presence of an object (e.g., object 110) is assumed if the levelof correlation exceeds a threshold for at least one sense circuit or atleast one pair of sense circuits of the plurality of sense circuits 104a, 104 b, . . . , 104 n.

In an example implementation or operations of an object detectioncircuit 100, inductive kinematic sensing is combined with aspace-differential detection scheme. Presence of an object (e.g., object110) is assumed if at least one difference between the correlator outputassociated to at least one first sense circuit (e.g., sense circuit 104a) and the correlator output associated to at least one second sensecircuit (e.g., sense circuit 104 b) exceeds a threshold. In anotherexample implementation or operation, correlator outputs obtained withinductive kinematic sensing, inductive thermal sensing, and inductiveferromagnetic sensing are combined.

In some implementations or operations of an object detection circuit100, one or more specialized detection schemes as described above areapplied initially before starting regular wireless power transfer e.g.,for the purpose of charging an electric vehicle. If the object detectioncircuit 100 determines presence of an object (e.g., object 110), thewireless power transfer system 200 stops detection based on aspecialized detection scheme and does not start regular wireless powertransfer. During regular wireless power transfer, the object detectioncircuit 100 may employ at least one other detection scheme e.g., atime-differential scheme capable of detecting an object (e.g., object110) when it enters the predetermined space.

In other implementations or operations of an object detection circuit100, one or more specialized detection schemes as described above areapplied in the event that the object detection circuit 100 trips duringregular wireless power transfer. In this case, one or more specializeddetection schemes may be used to verify this detection by first ceasingregular power transfer and then reapplying one or more specializeddetection schemes. This may apply in case of a detection with lowconfidence. If the previous finding of presence of an object (e.g.,object 110) is confirmed, the wireless power transfer system 200discontinues regular wireless power transfer else it reactivates regularwireless power transfer.

In some aspects of an object detection circuit 100, the control circuit950 and/or the control, processing & evaluation circuit 960 areconfigured to perform calibration for purposes of determining referencevalues as previously described for purposes of threshold detection. Infurther aspects, they may be configured to perform calibration of theimpedance or transimpedance measurement with respect to the angle (e.g.,arg{Z₁}) as previously described with reference to the circuit 300 ofFIG. 3A.

In other aspects of an object detection circuit 100, the control circuit950 and/or the control, processing & evaluation circuit 960 areconfigured to perform functions such as finding optimum sensefrequencies f_(s) minimizing the effect of circuit extrinsic noise aspreviously described with reference to the circuit 300 of FIG. 3A.

In some implementations or operations of an object detection circuit100, the driver circuit 910 is configured to concurrently generate andselectively (e.g., sequentially) apply a plurality of current signals I₁each with a distinct frequency f_(s). Each of the plurality of currentsignals I₁ is selectively (e.g., sequentially) applied to each of asubset of the plurality of sense circuits 104 a, 104 b, . . . , 104 n.The number of subsets may equal the number of concurrently appliedcurrent signals I₁. As previously mentioned, such multiple currentsignal operation may be used to expedite impedance and/or transimpedancemeasurements in each of a plurality of sense circuits and/or in each ofa plurality of pairs of sense circuits, respectively, of the pluralityof sense circuits 104 a, 104 b, . . . , 104 n.

In some implementations or operations where two or more current signalsI₁ are applied at a time, intermodulation effects may occur e.g., due tocross-talk in multiplexers 916 and 922, in the plurality of sensecircuits 104 a, 104 b, . . . , 104 n, and residual non-linearity e.g.,in the measurement circuit 920. If three or more current signals I₁ eachwith a distinct frequency f_(s) are concurrently applied,intermodulation products may fall on one or more of the applied sensefrequencies f_(s) thus interfering with the one or more voltage signalsV₂ in the measurement circuit 920. In an example operation where threecurrent signals I₁ each with a distinct frequency f_(s) are concurrentlyapplied, an intermodulation product generated by the first and secondcurrent signal I₁ having a first and second frequency f_(s) may fallonto the third frequency f_(s) of the third current signal I₁, thusinterfering e.g., in the measurement circuit 920 with the voltage signalV₂ in response to the third current signal I₁. Interference caused byintermodulation products generally manifests in variations in a sequence(time-series) of consecutive measurement outputs V_(out) associated tothe same at least one sense circuit (e.g., sense circuit 104 a or pairof sense circuits 104 a and 104 b). If measurement outputs V_(out) referto an output (not shown in FIG. 9) in the control, processing &evaluation circuit 960, these variations may be generally characterizedby variations in a magnitude and phase. These variations may resemblethe effects of circuit intrinsic or extrinsic noise sources aspreviously discussed with reference to FIG. 3A. However, the variancecaused by intermodulation interference may be significantly larger thanthat produced e.g., by circuit intrinsic noise. These variations mayrequire the detection threshold to be increased e.g., in animplementation or operation based on a time-differential detectionscheme as described below, thus resulting in lower detectionsensitivity.

In some implementations or operations of an object detection circuit100, interference caused by intermodulation effects is mitigated by asmart selection of sense frequencies f_(s) e.g., by selecting a set offrequencies whose intermodulation products (e.g., up to the fourthorder) do not fall on any of the frequencies f_(s) of the selected setof frequencies. However, this approach may put some constraints on thechoice of sense frequencies f_(s). This may be true if intermodulationproducts of several orders need to be taken into account, the number ofconcurrently applied currents I₁ is greater than three and inimplementations or operations where sense circuits have to be operatedat or near resonance, have tolerances with respect to the resonantfrequency, and where some frequency bands e.g., jammed by frequencyselective switching noise have to be avoided for purposes as previouslydiscussed.

In another implementation or operation of an object detection circuit100, interference due to intermodulation effects is mitigated bychoosing sense frequencies f_(s) for the plurality of concurrentlyapplied current signals I₁ from a set of frequencies with a fixedfrequency spacing Δf (e.g., a frequency grid). In some implementations,each sense frequency f_(s) is a multiple of the frequency spacing Δf(e.g., 1 kHz). It may be appreciated that the plurality of currentsignals I₁ with sense frequencies f_(s) differing by integer multiplesof Δf generates intermodulation products with frequencies that are alsoan integer multiple of Δf, thus may potentially fall on one or moresense frequencies f_(s). Additionally, the phase of each of theplurality of concurrently applied current signals I₁ is controlled in amanner so that any intermodulation product falling on a sensefrequencies f_(s) produces a static offset (instead of noise-likevariations) in sequences (time-series) of consecutive measurementoutputs V_(out) associated to the same at least one sense circuit (e.g.,sense circuit 104 a or pair of sense circuits 104 a and 104 b). It maybe appreciated that a small static offset (e.g., equivalent to afractional change below 1%) as it may be caused by intermodulationeffects may not negatively impact operation of an object detectioncircuit 100. This may be particularly true for implementations oroperations employing a time-differential detection scheme.

In some implementations or operations of the object detection circuit100 of FIG. 1 employing multiple sense current signal operation andphase control to mitigate intermodulation effects, the phase of each ofthe plurality of sinusoidal current signals I₁ that is concurrently andselectively (e.g., sequentially) applied is reset (e.g., to zero) at thestart of a measurement time interval. More precisely, in a multiplesense current signal operation where each of a plurality of currentsignals I₁ is concurrently and sequentially applied to each sensecircuit of a subset of the plurality of sense circuits 104 a, 104 b, . .. , 104 n as previously described, the phase of each current signal I₁is reset at the start of each measurement time interval as provided forsequentially measuring an electrical voltage V₂ in each of the pluralityof sense circuits 104 a, 104 b, . . . , 104 n in a time-multiplexedfashion.

In some aspect of an object detection circuit 100, the control,processing & evaluation circuit 960 is also configured to extract aresidual low frequency signal component e.g., at the wireless powertransfer frequency f_(wpt) (e.g., 85 kHz) as it may exist in themeasurement output V_(out) of the at least one measurement amplifiercircuit 924. Such low frequency signal component may be present duringwireless power transfer or whenever the wireless power transferstructure 224 generates a low frequency magnetic field. To some extent,such low frequency signal component may be also present if thevehicle-side wireless power transfer structure 260 generates a lowfrequency magnetic field e.g., at the wireless power transfer frequencyf_(wpt). In implementations using capacitive compensation of animpedance or mutual impedance e.g., based on the circuit 400 of FIG. 4Aor on the circuit 800 of FIG. 8A, respectively, this low frequencycomponent in the measurement output V_(out) may be substantiallyattenuated as previously discussed with reference to FIG. 3A and FIG.8A, respectively. In some aspect, this low frequency signal component isindicative for the electrical voltage induced into the sense element(e.g., sense element 106 a) of the plurality of sense elements 106 a,106 b, . . . , 106 n by the low frequency magnetic field as present atthe location of this sense element. It is also indicative for the lowfrequency signal component in the electrical voltage V₂ at therespective sense circuit (e.g., sense circuit 104 a). More precisely, insome aspect, this low frequency signal component in the measurementoutput V_(out) may be proportional to the low frequency signal componentin the electrical voltage V₂ at the sense circuit (e.g., sense circuit104 a) that is selectively connected to the measurement amplifiercircuit 924 via multiplexer 922. In some implementations or operationsof an object detection circuit 100, the level (e.g., r.m.s. level) ofthis low frequency component in the measurement output V_(out) isproportional to the level of the low frequency magnetic flux passingthrough the sense element (e.g., sense element 106 a) associated to themeasurement output V_(out).

In some implementations or operations an object detection circuit 100,the control, processing & evaluation circuit 960 may use the lowfrequency component extracted from the measurement output V_(out) forvarious purposes e.g., in connection with the specialized detectionschemes as previously described.

In some implementations or operations, the control, processing &evaluation circuit 960 is configured to use the extracted low frequencycomponent being indicative for the level of the low frequency magneticfield at the location of the sense element (e.g., sense element 106 a)for correlation. More precisely, it is configured to correlate asequence (time-series) of low frequency components extracted fromconsecutively measured V_(out), each associated to the same at least onesense circuit (e.g., sense circuit 104 a or pair of sense circuits 104 aand 104 b) with a corresponding sequence of consecutively measuredV_(out) being indicative for the electrical voltage V₂ at the sensefrequency f_(s). In another example implementation or operationemploying inductive thermal sensing, the control, processing &evaluation circuit 960 is configured to correlate a sequence ofconsecutively measured low frequency components each associated to thesame at least one sense circuit (e.g., sense circuit 104 a or pair ofsense circuits 104 a and 104 b) with a corresponding sequence of a timederivative of consecutively measured V_(out) being indicative for theelectrical voltage V₂ at sense frequency f_(s). In a further exampleimplementation or operation employing inductive thermal sensing thecontrol, processing & evaluation circuit 960 is configured to use otherfunctions (e.g., similar to the time derivative) to modify the sequenceof measurement outputs V_(out) indicative for the electrical voltage V₂at sense frequency f_(s).

In other implementations or operations, the control, processing &evaluation circuit 960 is configured to use the extracted low frequencycomponent being indicative for the low frequency magnetic field signalat the location of the sense element (e.g., sense element 106 a) forcorrelation. In an example implementation or operation employinginductive ferromagnetic sensing, the control, processing & evaluationcircuit 960 is configured to correlate a sequence (time-series) of lowfrequency components, each associated to the same at least one sensecircuit (e.g., sense circuit 104 a or pair of sense circuits 104 a and104 b) with a corresponding sequence of consecutively measured V_(out)being indicative for the electrical voltage V₂ at sense frequency f_(s).In an example implementation or operation, the control, processing &evaluation circuit 960 is configured to modify the sequence of lowfrequency components prior correlation e.g., by applying an appropriatefunction so that the new modified sequence is representative for the lowfrequency magnetic field signal waveform as it would look like afterrectification.

In further implementations or operations, the control, processing &evaluation circuit 960 is configured to use the extracted low frequencycomponent as supplementary information being indicative of presence andlevel of the low frequency magnetic field at wireless power transferfrequency f_(wpt) e.g., to improve the evaluation of measurement outputsV_(out) to determine presence of an object (e.g., object 110). As anon-limiting example, this improvement may include adaptation of athreshold e.g., to reduce a false detection rate e.g., due to externalnoise at sense frequency f_(s) generated by the power conversion circuit222 with reference to FIG. 2A, which may accompany the low frequencymagnetic field.

In yet another implementation or operation, the control, processing &evaluation circuit 960 is configured to use the extracted low frequencycomponent as supplementary information for determining presence of anobject (e.g., object 110). This low frequency component may beindicative e.g., for the magnitude and phase of the magnetic flux thatis generated by the wireless power transfer coil 226 and that passesthrough a sense element (e.g., sense element 106 a that may include asense coil e.g., sense coil 302 with reference to FIG. 3A). It may beappreciated that the magnitude and phase of the magnetic flux and thusof the extracted low frequency component associated to a sense element(e.g., sense element 106 a) may change if an object (e.g., object 110)is in proximity of that sense coil. Therefore, in some implementationsor operations of a control, processing & evaluation circuit 960 isconfigured to additionally evaluate magnitude and phase of the extractedlow frequency component associated to each of the plurality of sensecircuits 104 a, 104 b, . . . , 104 n for determining presence of anobject (e.g., object 110) in proximity to the at least one of theplurality of sense elements 106 a, 106 b, . . . , 106 n. This sensingtechnique based on the extracted low frequency component in measurementoutputs V_(out) may be considered as measuring a change in atransimpedance (e.g., mutual impedance) between the wireless powertransfer coil 226 and each of the plurality of sense elements 106 a, 106b, . . . , 106 n.

As described above with reference to the measurement and detectioncircuit 108 of FIG. 1, at least a portion of the control circuit 950and/or the control, processing & evaluation circuit 960 may beimplemented by one or more micro-controllers or other type processors.This portion may be implemented as an application-specific integratedcircuit (ASIC), a field programmable gate array (FPGA) device, a digitalsignal processor (DSP), another processor device or combinationsthereof. The control circuit 950 and/or the control, processing &evaluation circuit 960 may be configured to receive information fromeach of the components of the circuit 900 and perform calculations basedon the received information. The control circuit 950 and/or the control,processing & evaluation circuit 960 may be configured to generatecontrol signals for each of the components that may adjust the operationof that component. The control circuit 950 and/or the control,processing & evaluation circuit 960 may further include a memory (notshown) configured to store data, for example, such as instructions forcausing the circuit 900 to perform particular functions, such as thoserelated to object detection. In some implementations, certain portionsof circuitry, components and/or processors of control circuit 950 andthe control, processing & evaluation circuit 960 may be shared orcombined. In addition, while different blocks are shown for purposes ofillustration, it is noted that the circuitry and/or components involvedin the driver circuit 910, the control circuit 950, the plurality ofsense circuits 104 a, 104 b, . . . , 104 n, the measurement circuit 920,and the control, processing & evaluation circuit 960 may be combined indifferent ways on one or more circuit boards or integrated circuitsregardless of any particular line shown separating different blocks.

The dashed lines used in FIG. 9 are to emphasize that the components andtheir configuration in the driver circuit 910 and the measurementcircuit 920 are illustrative and other implementations may have these orother components configured to selectively drive the plurality of sensecircuits 104 a, 104 b, . . . , 104 n with at least one sense currentsignal I₁ and to selectively measure at least one electrical voltage V₂in the plurality of sense circuits 104 a, 104 b, . . . , 104 n,respectively. Furthermore, while certain circuit elements are describedas connected between other elements it should be appreciated that theremay be other circuit elements in various implementations that may alsobe in between the two elements described as electrically connected(e.g., other elements interposed).

FIG. 10 is a circuit diagram of a circuit 1000 illustrating an exampleimplementation of a portion of an object detection circuit 100. Thecircuit 1000 of FIG. 10 illustrates the analog front-end circuit portionof the object detection circuit 100 of FIG. 1 and for purposes ofillustration may exclude various other signal generation, processing andevaluation circuitry (e.g., control circuit 950 and control, processing& evaluation circuit 960 with reference to FIG. 9) that may be needed insome implementations of an object detection circuit 100. As furtherexplained below, the circuit 1000 may apply to certain impedance andmutual impedance sensing techniques as previously described withreference to FIGS. 4A and 5A. The circuit 1000 is based on the currentsource voltage measurement approach as previously described inconnection with FIG. 3A and may be subdivided into a driver circuit 910,a plurality of sense circuits 104 a, 104 b, . . . , 104 n, and ameasurement circuit 920 as previously described with reference to thegeneric block diagram of FIG. 9.

In the exemplary implementation shown in FIG. 10, each of the pluralityof sense circuits 104 a, 104 b, . . . , 104 n have an identical circuittopology. Therefore, descriptions given below for the sense circuit 104a also apply to the other sense circuits (e.g., 104 b, . . . , 104 n).The sense circuit 104 a includes sense element 106 a including a sensecoil 302 (e.g., a planar multi-turn coil), a first capacitor 420, ashunt inductor 1008, a second capacitor 1006, and a third capacitor1010. The first capacitor 420 also referred to as the compensation ortuning capacitor is connected electrically in series with the senseelement 106 a forming a series-resonant circuit. If operated at a sensefrequency f_(s) near resonance, the first capacitor 420 compensates forthe gross portion of the reactance of the sense element 106 a (sensecoil 302) for purposes as previously discussed with reference to FIGS.4A and 4B.

In some implementations, capacitors 420, 1006, and 1010 may be of a typewith a low temperature coefficient providing high thermal stability(e.g., a NP0-type capacitor) reducing thermal drift of an electricalcharacteristic (e.g., an impedance) as measured at each of the pluralityof sense circuits 104 a, 104 b, . . . , 104 n.

Moreover, as previously mentioned in connection with the circuit 400 ofFIG. 4A, the first capacitor 420 may act as a high pass filter toattenuate the high voltages that may be induced into the sense coil 302by the strong magnetic fields associated with the wireless powertransfer at a frequency f_(wpt). Therefore, the first capacitor 420 mayalso serve for protecting the sense coil 302, the components of thedriver circuit 910 as well as the measurement circuit 920 e.g., fromdamage by excessive current flow, consequent heating effects,overloading, or surpassing some voltage limits. To more effectivelyattenuate any signal component at the wireless power transfer frequencyf_(wpt) and low frequency harmonics thereof in the electrical voltageV₂, a shunt inductor 1008 is connected in parallel to (across) theseries circuit of capacitor 420 and sense coil 302 as shown in FIG. 10.The capacitor 420 together with the shunt inductor 1008 form a 2^(nd)order high pass filter that is configured to attenuate these lowfrequency signal components to a level e.g., significantly below thelevel of the electrical voltage V₂ in response to a respective sensecurrent I₁ at sense frequency f_(s) (e.g., in the MHz range). Therefore,this high pass filter may substantially reduce dynamic rangerequirements in the measurement circuit 920 and in a further processing(e.g., ADC) e.g., as part of the control, processing & evaluationcircuit 960 of FIG. 9. It may also reduce any cross-modulation effectsbetween any low frequency signals at the wireless power transferfrequency f_(wpt) and harmonics thereof and the sense signal atfrequency f_(s). Cross-modulation may be produced by residual non-lineareffects in the measurement circuit 920. At sense frequencies f_(s), thishigh pass filter may exert a minor impact on the voltage (impedance)measurement and which may be corrected (compensated for) in a furtherprocessing (e.g., in the control, processing & evaluation circuit 960shown in FIG. 9). Any phase shift caused by this high pass filter may bedetermined e.g., by performing an impedance calibration with respect tothe angle e.g., arg{Z_(1c)} as previously mentioned with reference toFIGS. 3A and 9.

An exemplary sense circuit (e.g., sense circuit 104 a) designed for anominal sense frequency f_(s)=3 MHz may use a sense coil 302 with aninductance L₁=5 μH, a compensation capacitor 420 with a capacitanceC=560 pF, and a shunt inductor 1008 of L_(sh)=5 μH. In this example, forillustrative purposes only, the sense circuit may provide 60 dBattenuation of the voltage induced into sense coil 302 at a wirelesspower frequency f_(wpt)=85 kHz. Assuming an induced voltage V_(ind)=30 Vacross the sense coil's 302 terminals, the component at f_(wpt) in thevoltage V₂ across the sense circuit (e.g., sense circuit 104 a) may beonly 30 mV while the wanted signal component at sense frequency f_(s)may be in the order of 100 mV.

The second capacitor 1006 may be needed in some implementations to blockany residual DC flow at the output of the driver circuit 910 (e.g., dueto a DC offset). In some aspect, the capacitor 1006 may also help toattenuate any residual low frequency current component (e.g., at awireless power frequency f_(wpt)) at the output of the driver circuit910. Moreover, in some implementations, it may be also used tocompensate or partially compensate for the effect of the reactance ofshunt inductor 1008 in the measured impedance Z_(1c)=V₂/I₁. Likewise,the third capacitor 1010 may be needed in some implementations to blockany residual DC flow at the input of the measurement circuit 920 (e.g.,due to a DC offset). In some aspect, the capacitor 1010 may also help toattenuate any residual low frequency current component (e.g., at awireless power frequency f_(wpt)) at the input of the measurementcircuit 920. Moreover, in some implementations, it may be also used tocompensate or partially compensate for the effect of the reactance ofshunt inductor 1008 in the measured impedance Z_(1c).

The driver circuit 910 includes a signal source 912, a driver amplifiercircuit 914, an input multiplexer circuit 916 illustrated in FIG. 10 asa plurality of switches 1016 a, 1016 b, . . . , 1016 n, and a pluralityof series resistors 1018 each having a resistance R_(ser1). The drivercircuit 910 is configured to operate as a current source (e.g., currentsource 306 as described in connection with FIG. 3A) and selectively(e.g., sequentially) apply a sense current signal I₁ at an operating(sense) frequency f_(s) to each of the plurality of sense circuits 104a, 104 b, . . . , 104 n. The signal source 912 generates the sensesignal e.g., a sinusoidal sense signal with a defined frequency f_(s),amplitude and phase. The driver amplifier circuit 914 as illustrated inFIG. 10 includes an operational amplifier 1015 (e.g., a low noiseoperational amplifier) and is configured to receive the sense signalfrom signal source 912 and to provide a corresponding voltage sourceoutput with an output voltage sufficient to selectively (e.g.,sequentially) drive each of the plurality of sense circuits 104 a, 104b, . . . , 104 n with a specified sense current I₁ at a sense frequencyf_(s). The driver amplifier circuit 914 together with series resistors1018 mimic a current source characteristic at each of the plurality ofoutputs of the driver circuit 910. It may be appreciated that seriesresistor 1018 with a large enough resistance R_(ser1) potentiallytransforms the voltage source output (virtually zero source impedance)of the driver amplifier circuit 914 into a quasi-ideal current source306 (high source impedance) with reference to FIG. 3A. Given an outputvoltage constraint for the driver amplifier circuit 914, a seriesresistor 1018 with a very large resistance R_(ser1) may provide analmost ideal current source characteristic but may also result in a toolow sense current I₁ and thus in insufficient SNR as it may be seen fromEquation (58). Therefore, in some implementations, the resistanceR_(ser1) may be a trade-off between the quality of the current source(e.g., as specified with reference to FIG. 3A) and the resulting SNR. Inan example implementation using an operational amplifier 1015 with anoutput voltage constraint of 3 Vrms, the trade-off may be found in aresistance R_(ser1)=100Ω resulting in a short circuit output currentI₁=30 mA. The quality of this current source may be assessed using thedefinition of Equation (inequality) (2) assuming a sense frequencyf_(s)=3 MHz, an example sense circuit (e.g., sense circuit 106 a)operated at resonance with an equivalent loss resistance (seriesresonant resistance) R₁=3Ω. The following expression may be found forthe left side of Equation (2):

$\begin{matrix}{{{{{\frac{\Delta \; V_{2}}{V_{2,0}}}{\frac{I_{1,0}}{\Delta \; I_{1}}}} \cong \frac{R_{{ser}\; 1} + R_{1}}{R_{1}}} = {34.3 > 10}},} & (82)\end{matrix}$

where |ΔV₂/V_(2,0)| refers to the fractional change in the voltage V₂e.g., at the sense circuit 104 a caused by an object (e.g., object 110)and |ΔI₁/I_(1,0)| to the resulting fractional change in the respectivecurrent I₁. The numerical result of Equation (82) indicates that thisexample implementation of a driver circuit 910 satisfies therequirements specified by Equation (2).

Within limits of the operational amplifier 1015, the driver circuit 910has the potential to drive more than one sense circuit with a current I₁at a time, each having the same frequency f_(s) and substantially thesame amplitude and phase. If e.g., switch 1016 a and 1016 b are closed,each of the sense circuits 104 a and 104 b is driven with a current I₁.

The input multiplexer circuit 916 includes a plurality of switches 1016a, 1016 b, . . . , 1016 n and is configured to selectively connect eachof the plurality of sense circuits 104 a, 104 b, . . . , 104 n viaseries resistor 1018 to the driver amplifier circuit 914 to selectively(e.g., sequentially) drive each of the plurality of sense circuits 104a, 104 b, . . . , 104 n with a sense current I₁ at a sense frequencyf_(s). Therefore, each of the plurality of switches 1016 a, 1016 b, . .. , 1016 n is electrically connected to a common input node that iselectrically connected to the output of the driver amplifier circuit914. Each of the plurality of switches 1016 a, 1016 b, . . . , 1016 nmay be one of a semiconductor analog switch (e.g., a single FET switch,a complementary FET switch composed of a p-channel and a n-channel typeFET), a micro-mechanical (MEMS) switch or any other type of switchproviding a sufficiently high current signal attenuation in theoff-state. An analog switch (e.g., a complementary FET switch) may bemodelled by an on-state series resistance R_(sw), an off-state seriescapacitance C_(sw), a first and a second parasitic shunt capacitanceC_(sh1) and C_(sh1), respectively, as indicated in FIG. 10 by the dashedline symbols for the switch 1016 b by example. It may be appreciatedthat the on-state series resistance R_(sw) (e.g., of switch 1016 a) maybe uncritical for the functioning of the driver circuit 910 since itmerges into an overall series resistance R_(ser1). Semiconductorswitches may be subject of temperature variations larger than those ofnormal resistors (e.g., metal film resistors). However, ifR_(sw)<<R_(ser1) holds, the switch (e.g., switch 1016 a) may notsignificantly deteriorate temperature stability of the overallresistance R_(ser1). It may also be appreciated that the totalcapacitive load produced by the plurality of first shunt capacitancesC_(sh1) may not noticeably impair the current source characteristic ofthe driver circuit 910 since it is in parallel to the voltage sourceoutput (virtually zero source impedance) of the driver amplifier circuit914. This may be true as long as the operational amplifier 1015 canhandle the resulting total capacitive load. Assuming operations whereonly one or a few switches of the plurality of switches 1016 a, 1016 b,1016 n are closed at a time, the total capacitive load at the commoninput node is governed by the plurality of first capacitances C_(sh1).Only one or a few second shunt capacitances C_(sh2) will add to thetotal capacitive load via the switch on-state series resistance R_(sw).An exemplary multiplexer circuit 916 for a sense frequency f_(s)=3 MHzmay use complementary FET switches with an on-state resistanceR_(sw)=5Ω, an off-state series capacitance C_(sw)=3 pF (corresponding toa series reactance of 17.7 kΩ), a first shunt capacitance C_(sh1)=12 pF,and a second shunt capacitance C_(sh2)=12 pF.

The measurement circuit 920 includes a plurality of series resistors1021 each having a resistance R_(ser2), an output multiplexer circuit922 illustrated in FIG. 10 as a plurality of switches 1023 a, 1023 b, .. . , 1023 n and a measurement amplifier circuit 924. The measurementcircuit 920 is configured to selectively (e.g., sequentially) measure anelectrical voltage V₂ in each of the plurality of sense circuits 104 a,104 b, . . . , 104 n and to provide a measurement output V_(out)proportional to the respective electrical voltage V₂ at a level suitablefor further processing e.g., in the control, processing & evaluationcircuit 960 with reference to FIG. 9. The measurement amplifier circuit924 as illustrated in FIG. 10 includes an operational amplifier 1025(e.g., a low noise operational amplifier) and is configured to operateas a transimpedance amplifier using feedback resistor 1027. This circuittransforms an input current I_(in) into an output voltage V_(out)proportional to I_(in). Since the voltage at the negative input ofoperational amplifier 1025 is virtually zero (virtual ground), themeasurement amplifier circuit 924 exhibits a virtually zero inputimpedance. The measurement amplifier circuit 924 together with seriesresistors 1021 mimic a quasi-ideal voltage measurement circuit 304 withreference to FIG. 3A to selectively (e.g., sequentially) measure theelectrical voltage V₂ at each of the plurality of sense circuits 104 a,104 b, . . . , 104 n. They form together a sign inverting voltageamplifier with an output V_(out) proportional to the sign inverse of thevoltage V₂. It may be appreciated that series resistor 1021 with a largeenough resistance R_(ser2) potentially transforms the quasi-zeroimpedance input of the measurement amplifier circuit 914 into a highimpedance input as required for a quasi-ideal voltage measurementcircuit 304 with reference to FIG. 3A. A series resistor 1021 with avery large resistance R_(ser2) may provide an almost ideal measurementcircuit but may also result in a lower current I_(in) and thus ininsufficient SNR with respect to the measurement circuit's 920 intrinsicnoise. Increasing resistance R_(ser2) also requires the feedbackresistance R_(f) to be increased if a specified voltage gain of themeasurement circuit 960 needs to be met. However, a too high resistanceR_(f) may cause the measurement amplifier circuit 924 to becomeunstable. Therefore, in some implementations of a measurement circuit920, the resistance R_(ser2) may be a trade-off between the quality ofthe voltage measurement (as defined with reference to FIG. 3A),resulting SNR, and stability conditions. In an example implementation,the measurement circuit 920 is designed to provide an output voltageV_(out)=350 mVrms when the sense circuit (e.g., sense circuit 104 a) isoperated at resonance and driven with a current I₁=30 mA. An equivalentloss resistance (resonance resistance) R₁=3Ω is assumed for each of theplurality sense circuits 104 a, 104 b, . . . , 104 n. In this exampleimplementation, the trade-off may be found in a series resistor 1012with a resistance R_(ser2)=150 S and a feedback resistor with aresistance R_(f)=590Ω resulting in a voltage gain of the measurementcircuit 920 of about 4. The quality of this quasi-ideal voltagemeasurement circuit may be assessed using the definition of Equation(inequality) (3) assuming resonance conditions and a resonant resistanceR₁=3Ω. The following expression may be found for the left side ofEquation (3):

$\begin{matrix}{{\frac{ɛ}{V_{{out},\infty}} = {\frac{{V_{out} - V_{{out},\infty}}}{V_{{out},\infty}} = {\frac{R_{1}}{R_{{ser}\; 2}} = {0.02 < 0.1}}}},} & (83)\end{matrix}$

where V_(out) refers to the measurement output of the measurementcircuit 920 (quasi-ideal voltage measurement circuit with a finite inputimpedance) and V_(out,∞) to the measurement output of an ideal voltagemeasurement circuit (infinite input impedance) but the same voltagegain. The numerical result of Equation (83) indicates that this exampleimplementation of a measurement circuit 920 satisfies the requirementsspecified by Equation (3).

Within limits of the operational amplifier 1025, the measurement circuit920 has the potential to measure the sum voltage of two or moreelectrical voltages V₂ at a time since it is configured as a summationamplifier. Series resistor 1021 transforms the voltage V₂ into acorresponding current at the common output node (input of themeasurement amplifier circuit 924). If e.g., switch 1023 a and switch1023 b are closed, the measurement output V_(out) is proportional to thesign inverse of the sum of the voltage V₂ at sense circuit 104 a andsense circuit 104 b, since currents V₂/R_(ser2) resulting at the commonoutput node add up.

The output multiplexer circuit 922 including the plurality of switches1023 a, 1023 b, . . . , 1023 n is configured to selectively connect eachof the plurality of sense circuits 104 a, 104 b, . . . , 104 n viaseries resistor 1021 to the measurement (transimpedance) amplifiercircuit 924 to selectively (e.g., sequentially) measure an electricalvoltage V₂ at each of the plurality of sense circuits 104 a, 104 b, . .. , 104 n in response to a current I₁ at a sense frequency f_(s).Therefore, each of the plurality of switches 1023 a, 1023 b, . . . ,1023 n is electrically connected to a common output node that iselectrically connected to the input of the measurement amplifier circuit924. As for the input multiplexer 916, each of the plurality of switches1023 a, 1023 b, . . . , 1023 n may be one of a semiconductor analogswitch (e.g., a single FET switch, a complementary FET switch composedof a p-channel and a n-channel type FET), a micro-mechanical (MEMS)switch or any other type of switch providing a sufficiently high currentsignal attenuation in the off-state. As previously described inconnection with the input multiplexer 916, each of the plurality ofswitches 1023 a, 1023 b, . . . , 1023 n may exhibit an on-state seriesresistance R_(sw), an off-state series capacitance C_(sw), a first and asecond parasitic shunt capacitance C_(sh1) and C_(sh1), respectively, asindicated in FIG. 10 by the dashed line symbols for the switch 1016 b byexample. It may be appreciated that the on-state series resistanceR_(sw) (e.g., of switch 1023 a) may be non-critical for the functioningof the measurement circuit 920 since it merges into an overall seriesresistance R_(ser2). If R_(sw)<<R_(ser2) holds, the switch 1023 b maynot significantly deteriorate temperature stability of the overallresistance R_(ser2) for reasons as previously discussed. It may also beappreciated that the total capacitive load produced by the plurality offirst parasitic shunt capacitances C_(sw1) at the common output node maynot impair the measurement circuit 920 since it is in parallel to thevirtually zero-impedance input of the measurement amplifier 924. Thismay be true as long as the operational amplifier 1025 can handle theresulting total capacitive load at its negative input. Assumingoperations where only one or a few switches of the plurality of switches1023 a, 1023 b, . . . , 1023 n are closed at a time, the totalcapacitive load is governed by the plurality of first capacitancesC_(sh1). Only one or a few second shunt capacitances C_(sh2) will add tothe total capacitive load via the switch' on-state resistance R_(sw). Anexemplary output multiplexer circuit 922 for a sense frequency f_(s)=3MHz may use complementary FET switches with the same characteristics asthose previously described for an example input multiplexer 916.

As previously mentioned, the circuit 1000 of FIG. 10 may apply tocertain impedance and mutual impedance sensing techniques e.g., aspreviously described with reference to FIGS. 4A and 5A. In an operationof an object detection circuit 100, the circuit 1000 is controlled in amanner to perform capacitively compensated impedance sensing withreference to the circuit 400 of FIG. 4A by selectively (e.g.,sequentially) applying a current I₁ to each of the plurality of sensecircuits 104 a, 104 b, . . . , 104 n and by selectively measuring anelectrical voltage V₂ at the corresponding sense circuit in response tothe current I₁ in a time-sharing mode. In some operations, switches ofthe input multiplexer 916 (plurality of switches 1016 a, 1016 b, . . . ,1016 n) and corresponding switches of the output multiplexer 922(plurality of switches 1023 a, 1023 b, . . . , 1023 n) are closedsequentially in a round robin fashion. In a first time interval, a firstswitch (e.g., switch 1016 a) of the input multiplexer 916 and acorresponding switch of the output multiplexer 922 (e.g., switch 1023 a)are closed and all other switches are open. In a second time interval, asecond input multiplexer switch (e.g., switch 1016 b) and acorresponding output multiplexer switch (e.g., switch 1023 b) are closedand all other switches are open. In a third time interval, a third inputmultiplexer switch (e.g., switch 1016 n) and a corresponding outputmultiplexer switch (e.g., switch 1023 n) are closed and all otherswitches are open, etc.

In another operation of an object detection circuit 100, the circuit1000 is controlled in a manner to perform mutual impedance sensing withreference to FIGS. 5A and 5D by selectively (e.g., sequentially)applying a current I₁ to each of the plurality of sense circuits 104 a,104 b, . . . , 104 n and by selectively (e.g., sequentially) measuringat least one electrical voltage V₂ at a different (non-corresponding)sense circuit of the plurality of sense circuits 104 a, 104 b, . . . ,104 n where no current I₁ is applied. In an example operation, eachswitch of the input multiplexer 916 (plurality of switches 1016 a, 1016b, . . . , 1016 n) and a different (non-corresponding) switch of theoutput multiplexer 922 (plurality of switches 1023 a, 1023 b, . . . ,1023 n) are closed sequentially in a round robin fashion. In a firsttime interval, a first input multiplexer switch (e.g., switch 1016 a)and a non-corresponding output multiplexer switch (e.g., switch 1023 b)are closed and all other switches are open. In a second time interval, asecond input multiplexer switch (e.g., switch 1016 b) and anon-corresponding output multiplexer switch (e.g., switch 1023 n) areclosed and all other switches are open. In a third time interval, athird input multiplexer switch (e.g., switch 1016 n) and anon-corresponding output multiplexer switch are closed and all otherswitches are open, etc. In this example operation of mutual impedancesensing, the pair of switches that are closed at the same time (e.g.,switch 1016 a and switch 1023 b) may be associated to a pair ofneighboring sense coils e.g., sense coil 106 a and 106 b of the sensecoil array 106. In this example operation of mutual impedance sensing,the pair of switches that are closed at the same time (e.g., switches1016 a and 1023 b) may be associated to a pair of neighboring sensecoils e.g., sense coil 106 a and 106 b of the sense coil array 106. Inanother example operation, some pairs of switches that are closed at thesame time (e.g., switches 1016 a and 1023 b) may be associated to pairsof neighboring sense coils (e.g., sense coils 106 a and 106 b), whileother switch pairs (e.g., switch 1016 b and switch 1023 n) may beassociated to non-neighboring sense coils (e.g., sense coils 106 b and106 n).

In some implementations based on the circuit 1000 of FIG. 10, the objectdetection circuit 100 is configured to perform impedance and mutualimpedance measurements with reference to FIGS. 4A, 5A and 5D forpurposes as previously described with reference to FIG. 9. In oneoperational example, it performs an impedance measurement for each ofthe plurality (N) of sense circuits 104 a, 104 b, . . . , 104 n and itperforms a mutual impedance measurement for each of at least a portion(subset) of the plurality of all possible pairs (N_(trans) combinationswith reference to Equation (80)) of sense circuits of the plurality ofsense circuits 104 a, 104 b, . . . , 104 n to determine whether anobject (e.g., object 110) is in proximity of at least one of a sensecoil of the plurality of sense coils 106 a, 106 b, . . . , 106 n.

FIG. 11 is a circuit diagram of a circuit 1100 illustrating an exampleimplementation of a portion of an object detection circuit 100. Similarto the circuit 1000 of FIG. 10, the circuit 1100 of FIG. 11 illustratesthe analog front-end circuit portion of the object detection circuit 100of FIG. 1 and for purposes of illustration may exclude various othersignal generation, processing and evaluation circuitry. As furtherexplained below, the circuit 1100 may apply to certain impedance andmutual impedance sensing techniques as previously described withreference to FIGS. 4A, 5A, and 7A. The circuit 1100 is based on thecurrent source voltage measurement approach as previously described inconnection with FIG. 3A and may be subdivided into a driver circuit 910,a plurality of sense circuits 104 a, 104 b, . . . , 104 n, and ameasurement circuit 920 as previously described with reference to theblock diagram of FIG. 9.

The circuit 1100 of FIG. 11 differs from the circuit 1000 of FIG. 10 bythe driver circuit 910. The driver circuit 910 is configured to operateas a dual current source to selectively (e.g., sequentially) apply twosense current signals I₁ at a time to each of the plurality of sensecircuits 104 a, 104 b, . . . , 104 n. A dual current source is requiredin an implementation or operation of a flux compensated mutual impedancesensing technique as illustrated by FIGS. 7A and 7D or 7E. Therefore,the driver circuit 910 includes a first and second signal source 912, afirst and second driver amplifier circuit 914, a first and second inputmultiplexer 916, and a first and second plurality of series resistors1018. For purposes of measuring an impedance or a mutual impedance, eachof the first and second signal source 912 may generate a sinusoidalsignal at a common sense frequency f_(s), but with a generally differentamplitude and phase as required to compensate for the magnetic flux in asense coil (e.g., sense coil 514) where the electrical voltage V₂ ismeasured and nullified as previously described with reference to FIG.7A. Each of the first and second driver amplifier circuit 914 mayinclude an identical operational amplifier 1015 (e.g., a low noiseoperational amplifier) and is configured to receive the sense signalfrom the respective signal source 912 and to provide a correspondingvoltage source output with an output voltage sufficient to selectively(e.g., sequentially) drive each of the plurality of sense circuits 104a, 104 b, . . . , 104 n with a specified sense current I₁ at a sensefrequency f_(s). The first input multiplexer 916 includes a firstplurality of switches 1016 a, 1016 b, . . . , 1016 n each electricallyconnected to the first driver amplifier circuit 914 and to a seriesresistor of the first plurality of series resistors 1018. Likewise, thesecond input multiplexer 916 includes a second plurality of switches1117 a, 1117 b, . . . , 1117 n each electrically connected to the seconddriver amplifier circuit 914 and to a series resistor of the secondplurality of series resistors 1018. As previously described withreference to FIG. 10, the first and second driver amplifier circuit 914together with series resistors 1018 mimic a current sourcecharacteristic at each of the first and second plurality of outputs ofthe driver circuit 910. For more detailed descriptions of the drivercircuit 910, the plurality of sense circuits 104 a, 104 b, . . . , 104n, and the measurement circuit 920, it is referred to the descriptionsof FIG. 10.

As previously mentioned, the circuit 1100 of FIG. 11 may apply tocertain impedance and mutual impedance sensing techniques e.g., aspreviously described with reference to FIGS. 4A, 5A, and 7A. In anoperation of an object detection circuit 1100, the circuit 1100 iscontrolled in a manner to perform flux compensated mutual impedancesensing with reference to the circuit 700 of FIG. 7A by selectively(e.g., sequentially) applying a first current I₁ to a first sensecircuit and a second current I₁ to a second sense circuit different fromthe first sense circuit of the plurality of sense circuits 104 a, 104 b,. . . , 104 n and by selectively measuring an electrical voltage V₂ at athird sense circuit different from the first and second sense circuit inresponse to the first and second current I₁ in a time-sharing mode. Inan operation based on the flux compensated mutual impedance sensingtechnique with reference to FIGS. 7A and 7D or 7E, a first switch of thefirst input multiplexer 916 (plurality of switches 1016 a, 1016 b, . . ., 1016 n), a non-corresponding second switch of the second inputmultiplexer 916 (plurality of switches 1117 a, 1117 b, . . . , 1117 n),and a non-corresponding third switch of the output multiplexer 1023 areclosed sequentially in a round robin fashion. The first, second, andthird switch are each associated to a different sense circuit of theplurality of sense circuits 104 a, 104 b, . . . , 104 n. In a first timeinterval, a first input multiplexer switch (e.g., switch 1016 a), asecond input multiplexer switch (e.g., switch 1117 b), and a thirdoutput multiplexer switch (e.g., switch 1023 n) are closed and all otherswitches are open. In a second time interval, a first input multiplexerswitch (e.g., switch 1016 b), a second input multiplexer switch (e.g.,switch 1117 n) (not shown in FIG. 11), and a third output multiplexerswitch (e.g., switch 1023 a) are closed and all other switches are open.In a third time interval, another combination of a first inputmultiplexer switch, a second non-corresponding input multiplexer switch,and a third non-corresponding output multiplexer switch are closed andall other switches are open, etc. In this example operation of fluxcompensated mutual impedance sensing, the pair of switches that areclosed at the same time (e.g., switches 1016 a, 1117 b, and 1023 n) maybe associated to a triple of neighboring sense coils e.g., sense coils106 a, 106 b, and 106 n, respectively, of the sense coil array 106. Inanother example operation, some triples of switches that are closed atthe same time (e.g., switches 1016 a, 1117 b, and 1023 n) may beassociated to a triple of neighboring sense coils (e.g., sense coils 106a, 106 b, and 106 n), while other triples may be associated tonon-neighboring sense coils.

In some implementations based on the circuit 1100 of FIG. 11, the objectdetection circuit 100 is configured to perform impedance and mutualimpedance measurements with reference to FIGS. 4A, 5A, and 7A forpurposes as previously described with reference to FIG. 9. In oneoperational example, it performs an impedance measurement for each ofthe plurality (N) of sense circuits 104 a, 104 b, . . . , 104 n and itperforms a mutual impedance measurement for each of at least a portion(subset) of the plurality of all possible triples (N_(fcmi) combinationswith reference to Equation (81)) of sense circuits of the plurality ofsense circuits 104 a, 104 b, . . . , 104 n to determine whether anobject (e.g., object 110) is in proximity of at least one of a sensecoil of the plurality of sense coils 106 a, 106 b, . . . , 106 n.

FIG. 12A is a block diagram of a circuit 1200 illustrating exampleimplementations of a portion of the object detection circuit 100 of FIG.1 based on the capacitively compensated mutual impedance sensingtechnique as previously described with reference to FIG. 8A. The blockdiagram may also apply to other transimpedance (e.g., mutual impedance)sensing techniques e.g., with reference to FIGS. 5A and 5D and tovarious measurement approaches such as the current source voltagemeasurement approach as previously described in connection with FIG. 3A.The circuit 1200 as shown in FIG. 12A includes a driver circuit 910, aplurality of sense circuits 104 a, 104 b, . . . , 104 n, a measurementcircuit 920, and a control, processing & evaluation circuit 960 aspreviously described with reference to FIG. 9 and for purposes ofillustration may exclude other circuitry (e.g., control circuit 950 withreference to FIG. 9) and electrical connections between circuits thatmay be needed in some implementations of the object detection circuit100 of FIG. 1.

Each of the plurality of sense circuits 104 a, 104 b, . . . , 104 n isconfigured for capacitively compensated mutual impedance sensing and hasan identical circuit topology. Each sense circuit (e.g., sense circuit104 a) includes a sense element comprised of a primary sense coil 512and a secondary sense coil 514 (e.g., planar multi-turn coils), and acapacitor 820. Capacitor 820 may also be referred to as compensation ortuning capacitor. A first terminal of capacitor 820 is electricallyconnected to a second terminal of each of the two sense coils 512 and514, while its second terminal is connected to a circuit ground. Thefirst terminal of the primary sense coil 512 is electrically connectedto a respective output of the driver circuit 910 while the firstterminal of the secondary sense coil 512 is electrically connected to arespective input of the measurement circuit 920. The two sense coils 512and 514 together with capacitor 820 form a series-resonant circuit withrespect to the mutual impedance as previously shown and described withreference to FIGS. 8A, 8B, 8C. If operated at a sense frequency f_(s)near resonance, capacitor 820 compensates for the gross portion of thereactance in the mutual impedance between primary sense coil 512 andsecondary sense coil 514 for purposes as previously discussed withreference to FIG. 8A.

The driver circuit 910 is electrically connected to the plurality ofsense circuits 104 a, 104 b, . . . , 104 n. In some implementations, thedriver circuit 910 is configured to operate as a current source (e.g.,current source 306 as described in connection with FIG. 3A) andselectively (e.g., sequentially) apply a sense current signal I₁ (asindicated in FIG. 12A) at a sense frequency f_(s) to each of theplurality of sense circuits 104 a, 104 b, . . . , 104 n. In otherimplementations, the driver circuit 910 is configured to operate as avoltage source e.g., as described in connection with FIG. 3A andselectively (e.g., sequentially) apply a sense voltage signal V₁ (asindicated in FIG. 12A) at an operating (sense) frequency f_(s) to eachof the plurality of sense circuits 104 a, 104 b, . . . , 104 n. In yetother implementations, the driver circuit 910 is configured to operateas any source that may be characterized neither by a current source nora voltage source. In certain implementations or operations, the drivercircuit 910 is configured to generate and apply more than one current I₁or voltage V₁ at a time for purposes as previously discussed inconnection with FIG. 9. To accomplish selective driving of each of theplurality of sense circuits 104 a, 104 b, . . . , 104 n, the drivercircuit 910 may include components such as at least one signal source912 (e.g., generating the sinusoidal sense signal with a definedfrequency f_(s), amplitude and phase), at least one driver amplifiercircuit 914, and at least one input multiplexer circuit 916.

The measurement circuit 920 is electrically connected to each of theplurality of sense circuits 104 a, 104 b, . . . , 104 n. In someimplementations, the measurement circuit 920 is configured toselectively (e.g., sequentially) measure an electrical voltage V₂ (asindicated in FIG. 12A) in each of the plurality of sense circuits 104 a,104 b, . . . , 104 n in response to the current I₁ being applied by thedriver circuit 910 to the respective sense circuit and to provide ameasurement output V_(out) proportional to the electrical voltage V₂. Inother implementations, the measurement circuit 920 is configured toselectively (e.g., sequentially) measure an electrical current I₂ (asindicated in FIG. 12A) in each of the plurality of sense circuits 104 a,104 b, . . . , 104 n in response to the voltage V₁ being applied by thedriver circuit 910 to the respective sense circuit and to provide ameasurement output V_(out) proportional to the electrical current I₂. Incertain implementations or operations, the measurement circuit 920 isconfigured to measure more than one voltage V₂ or electrical current I₂at a time and to provide more than one respective measurement outputV_(out) for purposes as previously discussed in connection with FIG. 9.To accomplish selective measurement of an electrical voltage V₂ orelectrical current I₂ in each of the plurality of sense circuits 104 a,104 b, . . . , 104 n, the measurement circuit 920 may include componentssuch as at least one output multiplexer circuit 922 and at least onemeasurement amplifier circuit 924. In some implementations e.g., basedon the current source voltage measurement approach, the at least onemeasurement amplifier circuit 924 may be configured to provide asufficiently high input impedance as needed for a quasi-ideal voltagemeasurement e.g., as defined with reference to FIG. 3A and low noise inits measurement output V_(out). In other implementations e.g., based onthe voltage source current measurement approach, the at least onemeasurement amplifier circuit 924 may be configured to provide asufficiently low input impedance as needed for a quasi-ideal currentmeasurement e.g., as defined with reference to FIG. 3A and low noise inits measurement output V_(out). The at least one multiplexer circuit 922may be electrically connected between the input of the respectivemeasurement amplifier circuit 924 and the plurality of sense circuits104 a, 104 b, . . . , 104 n and is configured to selectively connecteach of the plurality of sense circuits 104 a, 104 b, . . . , 104 n tothe input of the respective measurement amplifier circuit 922.

The control, processing & evaluation circuit 960 is configured toprovide further signal processing and evaluation of the acquiredmeasurement data. As previously discussed in connection with FIG. 9, itmay be configured to provide analog-to-digital signal conversion,frequency selective filtering, synchronous detection, combining outputs,summation of outputs, averaging of outputs, scaling of outputs,correction of outputs, evaluation of sequences (time-series) and/orpatterns of outputs indicative of an impedance and/or transimpedance(e.g., mutual impedance), etc. and eventually to decide whether anobject (e.g., object 110 of FIG. 3A) is proximate to at least one senseelement of the plurality of sense elements 106 a, 106 b, . . . , 106 nbased on a change in a transimpedance (e.g., capacitively compensatedmutual impedance) and to output at least a detection hypothesis H_(out)as indicated in FIG. 12A.

FIG. 12B is a circuit diagram of a circuit 1210 illustrating an exampleimplementation of a portion of the circuit 1200 of FIG. 12A. Similar tothe circuit 1000 of FIG. 10 and the circuit 1100 of FIG. 11, the circuit1210 illustrates the analog front-end circuit portion of the objectdetection circuit 100 of FIG. 1 and for purposes of illustration mayexclude various other signal generation, processing and evaluationcircuitry. Similar to the circuit 1200 of FIG. 12A, the circuit 1210applies to the capacitively compensated mutual impedance sensingtechnique as previously described with reference to FIG. 8A but may alsoapply to other transimpedance sensing techniques e.g., with reference toFIGS. 5A and 5D. The circuit 1210 is based on the current source voltagemeasurement approach as previously described in connection with FIG. 3Aand includes the driver circuit 910, the plurality of sense circuits 104a, 104 b, . . . , 104 n, and the measurement circuit 920 as previouslydescribed with reference to the block diagram of FIGS. 9 and 12A.

Each sense circuit of the plurality of sense circuits 104 a, 104 b, . .. , 104 n is configured for capacitively compensated mutual impedancesensing and have an identical circuit topology. Therefore, descriptionsgiven below for the sense circuit 104 a also apply to the other sensecircuits (e.g., 104 b, . . . , 104 n).

The sense circuit 104 a includes a sense element comprised of a primarysense coil 512 and a secondary sense coil 514 (e.g., planar multi-turncoils), a first capacitor 820, a first shunt inductor 1206, a secondcapacitor 1006, a second shunt inductor 1008, and a third capacitor1010. The first capacitor 820 may also be referred to as thecompensation or tuning capacitor is electrically connected to a secondterminal of each of the two sense coils 512 and 514 forming aseries-resonant circuit with respect to the mutual impedance aspreviously described with reference to FIG. 8A. If operated at a sensefrequency f_(s) near resonance, the first capacitor 820 compensates forthe gross portion of the reactance in the mutual impedance betweenprimary sense coil 512 and secondary sense coil 514 for purposes aspreviously discussed with reference to FIG. 8A. The first terminal ofthe primary sense coil 512 is electrically connected to a respectiveoutput of the driver circuit 910 via the second capacitor 1006 while thefirst terminal of the secondary sense coil 512 is electrically connectedto a respective input of the measurement circuit 920 via the thirdcapacitor 1010.

In some implementations, capacitors 820, 1006, and 1010 may be of a typewith a low temperature coefficient providing high thermal stability(e.g., a NP0-type capacitor) reducing thermal drift of thetransimpedance (e.g., capacitively compensated mutual impedance) asmeasured in each of the plurality of sense circuits 104 a, 104 b, . . ., 104 n.

Moreover, as previously mentioned in connection with the circuit 800 ofFIG. 8A, the first capacitor 820 may act as a high pass filter toattenuate the high voltages that may be induced into the sense coils 512and 514 by the strong magnetic fields associated with the wireless powertransfer at a frequency f_(wpt). Therefore, the first capacitor 820 mayalso serve for protecting the sense coils 512 and 514, the components ofthe driver circuit 910 and the measurement circuit 920 e.g., from damageby excessive current flow, consequent heating effects, overloading, orsurpassing some voltage limits. To more effectively attenuate any signalcomponent at wireless power transfer frequency f_(wpt) and low frequencyharmonics thereof in the current I₁, a shunt inductor 1206 is connectedin parallel to (across) the series circuit of capacitor 820 and primarysense coil 512 as shown in FIG. 12. The capacitor 820 together with theshunt inductor 1206 form a 2^(nd) order high pass filter that ispotential to attenuate these low frequency signal components in thecurrent I₁ to a level e.g., significantly below the current level atsense frequency f_(s) (e.g., in the MHz range). Therefore, this highpass filter may protect the driver amplifier circuit 914 and themultiplexer circuit 916 and may also reduce any cross-modulation effectsbetween any low frequency signals at the wireless power transferfrequency f_(wpt) and harmonics thereof and the sense signal atfrequency f_(s). Cross-modulation may be produced by residual non-lineareffects in the driver circuit 910. Likewise, a shunt inductor 1008 isconnected in parallel to (across) the series circuit of capacitor 820and the secondary sense coil 514 as shown in FIG. 12B to moreeffectively attenuate any signal component at the wireless powertransfer frequency f_(wpt) and low frequency harmonics thereof in theelectrical voltage V₂ at the input of the voltage measurement amplifiercircuit 920. Therefore, this high pass filter may substantially reducedynamic range requirements in the measurement circuit 920 and in afurther processing (e.g., ADC) e.g., as part of the control, processing& evaluation circuit 960 with reference to FIG. 9. The shunt inductor1008 may also reduce any cross-modulation effects between any lowfrequency signals at the wireless power transfer frequency f_(wpt) andharmonics thereof and the sense signal at frequency f_(s) (e.g., in theMHz range). Cross-modulation may be produced by residual non-lineareffects in the measurement circuit 920. At sense frequencies f_(s), theprimary and secondary side high pass filters of the sense circuit 104 amay exert a minor impact on the voltage (transimpedance) measurement andwhich may be corrected (compensated for) in a further processing (e.g.,in the control, processing & evaluation circuit 960 with reference toFIG. 9). Any phase shift caused by these high pass filters may bedetermined e.g., by performing a transimpedance calibration with respectto the angle e.g., arg{Z_(12c)} as previously mentioned with referenceto FIGS. 8A and 9.

As previously discussed in connection with FIG. 10, the second capacitor1006 may be needed in some implementations to block any residual DC flowat the output of the driver circuit 910 (e.g., due to a DC offset). Insome aspects, the capacitor 1006 may also help to attenuate any residuallow frequency current component (e.g., at a wireless power frequencyf_(wpt) and low frequency harmonics thereof) at the output of the drivercircuit 910. Moreover, in some implementations, it may be also used tocompensate or partially compensate for the effect of the reactance ofshunt inductor 1206 in the measured transimpedance Z_(12c)=V₂/I₁.Likewise, the third capacitor 1010 may be needed in some implementationsto block any residual DC flow at the input of the measurement circuit920 (e.g., due to a DC offset). In some aspect, the capacitor 1010 mayalso help to attenuate any residual low frequency voltage component(e.g., at a wireless power frequency f_(wpt) and low frequency harmonicsthereof) at the input of the measurement circuit 920. Moreover, in someimplementations, it may be also used to compensate or partiallycompensate for the effect of the reactance of shunt inductor 1008 in themeasured transimpedance Z_(12c).

The exemplary implementations of the driver circuit 910 and themeasurement circuit 920 as shown in FIG. 12B are not further describedherein as they may be identical to the driver circuit 910 and themeasurement circuit 920 of FIG. 10.

In some implementations of a circuit 1210 of FIG. 12B, sense coil 512and 514 are tightly coupled as previously discussed with reference toFIG. 8A and FIG. 5F. In this special case, the mutual inductance M₁₂approaches the self-inductance L₁ or L₂ of the sense coil 512 or 514,respectively, as expressed by Equation (77). As a consequence, capacitor820 compensates for the mutual reactance between sense coil 512 and 514as well as for the self-reactance of the individual sense coils 512 and514 substantially at the same frequency. In other words, resonance withrespect to the mutual inductance M₁₂ occurs substantially at the samefrequency as resonance with respect to each of the self-inductances L₁and L₂. If tuned on resonance, the load impedance as presented by thesense circuit 104 a to the driver circuit 910 is low and substantiallyequals the loss resistance R₁ of the primary sense coil 512 as shown inthe equivalent circuit of FIG. 8B. For a given output voltage of thedriver amplifier circuit 914 and a given resistance R_(ser1) of seriesresistor 1018, a low load impedance (e.g., substantially equal to theloss resistance R₁ and R₁<<R_(ser1)) may be beneficial to satisfy therequirements of a quasi-ideal current source as evident from Equation(82). Likewise, if tuned on resonance, the source impedance that ispresented by the sense circuit 104 a to the measurement circuit 920 islow and substantially equals the loss resistance R₂ of the secondarysense coil 514 as shown in the equivalent circuit of FIG. 8B. For agiven configuration of the measurement amplifier circuit 924 and a givenresistance R_(ser2) of series resistor 1021, a low source impedance(e.g., substantially equal to the loss resistance R₂ and R₂<R_(ser2))may be beneficial to satisfy the requirements of a quasi-ideal voltagemeasurement circuit as evident from Equation (82) if R₁ is replaced byR₂.

An exemplary circuit 1210 designed for a nominal sense frequency f_(s)=3MHz may use a sense coil 512 and 514 with substantially equal inductanceL₁≅L₂=5 μH and loss resistance R₁≅R₂=3Ω, a compensation capacitor 820with a capacitance C=560 pF, and shunt inductors 1006 and 1010 with aninductance L_(sh)=5 μH, an operational amplifier 1015 with an outputvoltage constraint of 3 Vrms, a plurality of series resistors 1018 witha resistance R_(ser1)=100Ω, and a plurality of resistors 1021 with aresistance R_(ser1)=150Ω. This exemplary circuit 1200 may provide afigure of merit for the driver circuit 910 of 34.3 with reference toEquation (82) and a figure of merit for the measurement circuit 920 withreference to Equation (83) of 0.02 as for the circuit 1000 of FIG. 10.

FIG. 13A is a circuit diagram of an alternative implementation (circuit1300) of a portion of the object detection circuit 100 of FIG. 1 basedon the capacitively compensated mutual impedance sensing technique usingthe current source voltage measurement approach. Similar to the circuit1000 of FIG. 10, the circuit 1300 illustrates the analog front-endcircuit portion of the object detection circuit 100 of FIG. 1 and forpurposes of illustration may exclude various other signal generation,processing and evaluation circuitry. The circuit 1300 differs from thecircuit 1210 of FIG. 12B by the implementation of the driver circuit910, the plurality of sense circuits 104, and the measurement circuit920. It uses less components and may therefore provide a BOM costadvantage over the circuit 1210 of FIG. 12B as explained in more detailbelow.

Each sense circuit (e.g., sense circuit 104 a) of the plurality of sensecircuits 104 includes a sense element (e.g., sense element 106 a)comprised of a primary sense coil 512 and a secondary sense coil 514.Each sense circuit (e.g., sense circuit 104 a) includes a switch cell1421 including a first switch (e.g., switch 1016 a) and a second switch(e.g., switch 1023 a) of the input and output multiplexer 916 and 922(with reference to FIG. 9), respectively. As compared to the circuit 900of FIG. 9, the circuit 1300 of FIG. 13A considers multiplexers 916 and922 as part of the plurality of sense circuits 104. The plurality ofsense circuits 104 further includes a first capacitor 820 (compensationcapacitor), a second capacitor 1006, and a third capacitor 1010. Asopposed to the circuit 1200 of FIG. 12, first, second, and thirdcapacitors of the circuit 1300 of FIG. 13A are common to the pluralityof sense circuits 104. A first terminal of each of the plurality ofprimary sense coils 512 is electrically connected to a first terminal ofcapacitor 1006, while a first terminal of each of the plurality ofsecondary sense coils 512 is electrically connected to a first terminalof capacitor 1010. The second terminal of each of the plurality ofprimary sense coils 512 is electrically connected to a first terminal ofa respective switch (e.g., switch 1016 a) of the plurality of switches1016 a, 1016 b, . . . , 1016 n, while the second terminal of each of theplurality of secondary sense coils 514 is electrically connected to afirst terminal of a respective switch (e.g., switch 1023 a) of theplurality of switches 1023 a, 1023 b, . . . , 1023 n. The secondterminal of each of the plurality of switches 1016 a, 1016 b, . . . ,1016 n and the plurality of switches 1023 a, 1023 b, . . . , 1023 n iselectrically connected to the circuit ground. Moreover, the plurality ofsense circuits 104 provides an input port and output port, each having afirst and a second terminal. Input and output port are common to theplurality of sense circuits. As opposed to the circuit 1200 of FIG. 12,the input port and the output port are common to the plurality of sensecircuits 104 a, 104 b, . . . , 104 n. The input port may be alsoreferred to as the driver port and the output port to the measurementport. The first terminal of the input port corresponds to the secondterminal of capacitor 1006, while the first terminal of the output portcorresponds to the second terminal of capacitor 1010. The secondterminal of the input port and the second terminal of the output portcorrespond to a first terminal of capacitor 820, while the secondterminal of capacitor 820 is electrically connected to circuit ground.

The driver circuit 910 is electrically connected to the common inputport of the plurality of sense circuits 104. It includes a signal source912, a driver amplifier circuit 914, and a series resistors 1018 common.The driver circuit 910 is configured to operate as a current source(e.g., current source 306 as described in connection with FIG. 3A) andselectively (e.g., sequentially) apply a sense current signal I₁ at anoperating (sense) frequency f_(s) to each of the plurality of sensecircuits 104 a, 104 b, . . . , 104 n. The signal source 912 generatesthe sense signal e.g., a sinusoidal sense signal with a definedfrequency f_(s), amplitude and phase. The driver amplifier circuit 914as illustrated in FIG. 13A includes an operational amplifier 1015 (e.g.,a low noise operational amplifier) and a transformer 1318 (e.g., a baluntransformer), and is configured to receive the sense signal from signalsource 912 and to provide a corresponding voltage source output with anoutput voltage sufficient to selectively (e.g., sequentially) drive eachof the plurality of sense circuits 104 a, 104 b, . . . , 104 n with aspecified sense current I₁ at a sense frequency f_(s). The transformer1318 is configured to convert the unbalanced (asymmetrical) output ofthe operational amplifier 1015 to a balanced (symmetrical) output forselectively driving each of the plurality of sense circuits 104 a, 104b, . . . , 104 n including the common (compensation) capacitor 820. Insome implementations, the transformer is a 1:1 transformer. In otherimplementations, it transforms the voltage by a ratio 1:2 or by anyother ratio 1:n₁. The driver amplifier circuit 914 together with seriesresistors 1018 mimic a current source characteristic at the output ofthe driver circuit 910 as previously described in connection with FIG.10.

The measurement circuit 920 includes a series resistor 1021 and ameasurement amplifier circuit 924. The measurement circuit 920 isconfigured to selectively (e.g., sequentially) measure an electricalvoltage V₂ in each of the plurality of sense circuits 104 a, 104 b, . .. , 104 n including the common compensation capacitor 820 and to providea measurement output V_(out) proportional to the respective electricalvoltage V₂ at a level suitable for further processing e.g., in thecontrol, processing & evaluation circuit 960 with reference to FIG. 9.The measurement amplifier circuit 924 as illustrated in FIG. 10 includesan operational amplifier 1025 (e.g., a low noise operational amplifier)and a transformer 1328 (e.g., a balun transformer) and is configured tooperate as a transimpedance amplifier using feedback resistor 1027. Thetransformer 1328 is configured to convert the unbalanced (asymmetrical)input of the operational amplifier 1025 to a balanced (symmetrical)input at the measurement amplifier circuit for selectively measuring anelectrical voltage V₂ in each of the plurality of sense circuits 104 a,104 b, . . . , 104 n including the common (compensation) capacitor 820.In some implementations, the transformer is a 1:1 transformer. In otherimplementations, it transforms the voltage by a ratio 1:2 or by anyother ratio 1:n₂. As previously described in connection with FIG. 10,the measurement amplifier circuit 924 together with series resistors1021 mimic a quasi-ideal voltage measurement circuit 304 with referenceto FIG. 3A to selectively (e.g., sequentially) measure the electricalvoltage V₂ at each of the plurality of sense circuits 104 a, 104 b, . .. , 104 n including the common (compensation) capacitor 820. They formtogether a sign inverting voltage amplifier with an output V_(out)proportional to the sign inverse of the voltage V₂.

In an operation of the circuit 1300, both switches of each of theplurality of switch cells 1421 including switches 1016 a and 1023 a,1016 b and 1023 b, . . . , 1016 n and 1023 n are selectively (e.g.,sequentially) closed to apply a sense current signal I₁ to thecorresponding sense circuit (e.g., sense circuit 104 a) of the of theplurality of sense circuits 104 a, 104 b, . . . , 104 n including thecommon (compensation) capacitor 820. In a first time interval, a firstinput multiplexer switch (e.g., switch 1016 a) and a correspondingoutput multiplexer switch (e.g., switch 1023 a) are closed and all otherswitches are open. In a second time interval, a second input multiplexerswitch (e.g., switch 1016 b) and a corresponding output multiplexerswitch (e.g., switch 1023 b) are closed and all other switches are open.In a third time interval, a third input multiplexer switch (e.g., 1016n) and a corresponding output multiplexer switch (e.g., 1023 n) areclosed and all other switches are open, etc. For example, if theswitches 1016 a and 1023 a are closed and all other switches are open,the primary sense coil 512 of the sense element 106 a is driven with asense current I₁ and the electrical voltage V₂ as indicated in FIG. 13Aat the common output port of the plurality of the sense circuits 104corresponds to the sum of the voltage across the secondary sense coil514 of sense circuit 104 a and the voltage across compensation capacitor820. If additionally the sense frequency f_(s) is adjusted for resonancewith respect to the mutual impedance of the sense circuit 104 a, thevoltage across capacitor 820 substantially compensates for the voltageacross sense coil 514 so that the electrical voltage V₂ at the commonoutput port of the plurality of sense circuits 104 becomes a minimum. Inother words, the mutual reactance between sense coil 512 and 514 ofsense circuit 104 a is compensated for by the reactance of capacitor820. If switches 1016 b and 1023 b are closed and all other switches areopen and if the sense frequency f_(s) is adjusted for resonance withrespect to the mutual impedance of the sense circuit 104 b, the mutualreactance between sense coil 512 and 514 of the sense circuit 104 b iscompensated for by the reactance of capacitor 820 and the electricalvoltage V₂ at the common output port of the plurality of sense circuits104 becomes a minimum, etc.

In another operation of the circuit 1300, a sense current signal I₁ isselectively (e.g., sequentially) applied to each (e.g., sense circuit104 a) of the plurality of sense circuits 104 including the common(compensation) capacitor 820 and the electrical voltage V₂ isselectively measured in a sense circuit (e.g., sense circuit 104 b)different from the sense circuit where the current I₁ is applied. Inthis operation, the mutual impedance between a sense coil 512 of a firstsense circuit (e.g., 104 a) and a sense coil 514 of a second sensecircuit (e.g., 104 b) is measured for detecting presence of an object inproximity of at least one of the sense coils. In a first time intervalof an example operation based on cross-coupling between sense coils ofdifferent sense circuits, a first input multiplexer switches (e.g.,switch 1016 a) and a non-corresponding output multiplexer switch (e.g.,switch 1023 b) are closed and all other switches are open. In a secondtime interval, a second input multiplexer switch (e.g., switch 1016 b)and a non-corresponding output multiplexer switch (e.g., switch 1023 n)are closed and all other switches are open. In a third time interval, athird input multiplexer switch (e.g., switch 1016 n) and anon-corresponding output multiplexer switch are closed, and all otherswitches are open, etc. For example, if the switches 1016 a and 1023 bare closed and all other switches are open, the primary sense coil 512of the sense circuit 104 a is driven with a sense current I₁ and theelectrical voltage V₂ as indicated in FIG. 13A at the common output portof the plurality of the sense circuits 104 corresponds to the sum of thevoltage across the secondary sense coil 514 of the sense circuit 104 band the voltage across compensation capacitor 820. If additionally, thesense frequency f_(s) is adjusted for resonance with respect to themutual impedance between sense coil 512 of the sense circuit 104 a andsense coil 514 of the sense circuit 104 b, the voltage across capacitor820 substantially compensates for the voltage across sense coil 514 ofthe sense circuit 104 b so that the electrical voltage V₂ at the commonoutput port of the plurality of sense circuits 104 becomes a minimum. Inother words, the mutual reactance between sense coil 512 of the sensecircuit 104 a and sense coil 514 of the sense circuit 104 b iscompensated for by the reactance of capacitor 820. If switches 1016 band 1023 n are closed and all other switches are open and if the sensefrequency f_(s) is adjusted for resonance with respect to the mutualimpedance between sense coil 512 of the sense circuit 104 b and sensecoil 514 of the sense circuit 104 n, the mutual reactance between sensecoil 512 of the sense circuit 104 b and sense coil 514 of the sensecircuit 104 c is compensated for by the reactance of capacitor 820 andthe electrical voltage V₂ at the common output port of the plurality ofsense circuits 104 becomes a minimum, etc.

It may be appreciated that the first (compensation) capacitor 820 mayalso act as a high pass filter to attenuate the high voltages that maybe induced into the sense coils 512 and 514 by the strong magneticfields associated with the wireless power transfer at a frequencyf_(wpt). Therefore, the first capacitor 820 may also serve forprotecting the sense coils 512 and 514 of each of the plurality of senseelements 106 a, 106 b, . . . , 106 n, each of the plurality of themultiplexer switch cells 1421, components of the driver circuit 910 andthe measurement circuit 920 e.g., from damage by excessive current flow,consequent heating effects, overloading, or surpassing some voltagelimits. To more effectively attenuate any signal component at wirelesspower transfer frequency f_(wpt) and low frequency harmonics thereof inthe current I₁, the transformer 1318 of the driver circuit 910 may beconfigured to provide a relatively low main inductance that appears inparallel to (across) the series circuit of capacitor 820 and the primarysense coil 512 of the selected sense circuit (e.g., sense circuit 104a). The capacitor 820 together with the transformer's 1318 maininductance form a 2^(nd) order high pass filter that is potential toattenuate these low frequency signal components in the current I₁ to alevel e.g., significantly below the current level at sense frequencyf_(s) (e.g., in the MHz range). The circuit 1300 may provide similarattenuation to the circuit 1200 of FIG. 12 using capacitor 820 and shuntinductor 1206. Therefore, this high pass filter may protect the driveramplifier circuit 914 as well as the plurality of switches 1016 a, 1016b, . . . , 1016 n and may also reduce any cross-modulation effects aspreviously discussed in connection with FIG. 10. Likewise, transformer1328 of the measurement circuit 920 may be configured to provide arelatively low main inductance that appears in parallel to (across) theseries circuit of capacitor 820 and the secondary sense coil 514 of theselected sense circuit (e.g., sense circuit 104 a) to more effectivelyattenuate any signal component at the wireless power transfer frequencyf_(wpt) and low frequency harmonics thereof in the electrical voltage V₂at the input of the voltage measurement amplifier circuit 920. Thecircuit 1300 may provide similar attenuation to the circuit 1200 of FIG.12 using capacitor 820 and shunt inductor 1008. Therefore, this highpass filter may substantially reduce dynamic range requirements in themeasurement circuit 920 and in a further processing (e.g., ADC) e.g., aspart of the control, processing & evaluation circuit 960 with referenceto FIG. 9. The shunt inductor 1008 may also reduce any cross-modulationeffects between any low frequency signals at the wireless power transferfrequency f_(wpt) and harmonics thereof and the sense signal atfrequency f_(s) (e.g., in the MHz range) as previously discussed inconnection with FIG. 10. At sense frequencies f_(s), these high passfilters that are effective at the common input and output of theplurality of sense circuits 104 may exert a minor impact on thetransimpedance measurement and may be corrected (compensated for) in afurther processing (e.g., in the control, processing & evaluationcircuit 960 with reference to FIG. 9). Any phase shift caused by thesehigh pass filters may be determined e.g., by performing a transimpedancecalibration with respect to the angle e.g., arg{Z_(12c)} as previouslymentioned with reference to FIGS. 8A and 9.

As previously discussed in connection with FIG. 10, the second capacitor1006 may be needed in some implementations to block any residual DC flowat the output of the driver circuit 910 (e.g., due to a DC offset). Insome aspect, the capacitor 1006 may also help to attenuate any residuallow frequency current component (e.g., at a wireless power frequencyf_(wpt) and low frequency harmonics thereof) at the output of the drivercircuit 910. Moreover, in some implementations, it may be also used tocompensate or partially compensate for the effect of the reactance ofthe transformer's 1318 main inductance in the measured transimpedanceZ_(12c)=V₂/I₁. Likewise, the third capacitor 1010 may be needed in someimplementations to block any residual DC flow at the input of themeasurement circuit 920 (e.g., due to a DC offset). In some aspect, thecapacitor 1010 may also help to attenuate any residual low frequencyvoltage component (e.g., at a wireless power frequency f_(wpt) and lowfrequency harmonics thereof) at the input of the measurement circuit920. Moreover, in some implementations, the capacitor 1010 may be alsoused to compensate or partially compensate for the effect of thereactance of the transformer's 1328 main inductance in the measuredtransimpedance Z_(12c).

In some implementations, capacitors 820, 1006, and 1010 may be of a typewith a low temperature coefficient providing high thermal stability(e.g., a NP0-type capacitor) reducing thermal drift of thetransimpedance (e.g., capacitively compensated mutual impedance) asmeasured e.g., in each of the plurality of sense circuits 104 a, 104 b,. . . , 104 n including capacitor 820.

In an example implementation of the circuit 1300 of FIG. 13A designedfor a nominal sense frequency f_(s)=3 MHz, an operational amplifier 1015with an output voltage constraint of 3 Vrms, a series resistor 1018 witha resistance R_(ser1)=100Ω, a transformer 1318 with a turns ratio of1:n₁=1:1 and a main inductance of L_(m1)=5 μH, a tightly coupled primaryand secondary sense coil 512 and 514 (coupling factor |k ₁₂|>0.9) withan inductance L₁=L₂=5 μH, a series resistor 1012 with a resistanceR_(ser2)=150Ω, a compensation capacitor 820 with a capacitance C=560 pF,a transformer 1328 with a turns ratio of 1:n₁=1:1 and a main inductanceof L_(m2)=5 μH, and a measurement amplifier circuit with a feedbackresistor with a resistance R_(f)=590Ω may be used.

As opposed to the circuit 1200 of FIG. 12, capacitors 820, 1006, 1010,series resistors 1018 and 1328, and transformers 1318 and 1328(replacing shunt inductors 1206 and 1008, respectively) are common tothe plurality of sense circuits 104 a, 104 b, . . . , 104 n, which mayprovide a BOM cost advantage over the circuit 1200 of FIG. 12.

In some implementations, a switch cell 1421 includes a pair of FETs asillustrated by the detail view of FIG. 13B. Since one terminal of eachFET is electrically connected to ground as shown in FIG. 13A, it may beappreciated that the gate drive circuit (not shown in FIGS. 13A and 13B)may be less complex compared to a non-ground connected analog switch(e.g., a complementary FET switch) as needed for the circuit 1200 ofFIG. 12. It may be also appreciated that the on-state resistance of theFET switches (e.g., switch 1016 a and 1023 a) may be uncritical for thefunctioning of the driver circuit 910 and the measurement circuit 920since it merges into an overall series resistance R_(ser1) and R_(ser2),respectively.

FIG. 14 is a flow diagram illustrating an example of a method 1400 fordetecting objects. The method is described with reference to FIG. 12A,however the techniques described may be implemented using any of thepreviously described elements of the example environment, components, orcircuits. Reference to elements is made by example only and is notintended to limit the ways in which the techniques can be implemented.Operations of a method may be repeated, combined, separated, omitted,performed in alternate orders, performed concurrently, or used inconjunction other methods illustrated in FIG. 14. At operational block1402, the method 1400 includes applying, from a driver circuit 910, aninput signal at an operating frequency to a first terminal of a primarysense coil 512, a second terminal of the primary sense coil 512electrically connected to a first terminal of a capacitor 820. Atoperational block 1404, the method 1400 further includes measuring, at ameasurement circuit 920, an electrical characteristic at an outputelectrically connected to a first terminal of a secondary sense coil514, a second terminal of the secondary sense coil electricallyconnected to the first terminal of the capacitor 820. At operationalblock 1406, the method 1400 further includes detecting whether an objectis in proximity to the primary or the secondary sense coil 512 and 514based on the electrical characteristic.

As described above the implementations of the object detection circuit100 of FIG. 1 described above may be implemented in the context of aninductive wireless power transfer system, and in particular a wirelesselectrical vehicle charging system where it may be desirable to detectobjects that may experience inductive heating due to the magnetic fieldused for power transfer.

FIG. 15 illustrates a wireless charging system 1500 for charging anelectric vehicle in which any of the elements/functionality describedwith reference to FIGS. 1-14 may be implemented.

The wireless charging system 1500 enables charging of an electricvehicle 1512 while the electric vehicle 1512 is parked so as towirelessly couple power from a base wireless power transfer unit 1502 a.The base wireless power transfer unit 1502 a may correspond to thewireless power transfer system 200 as described with reference to FIG.2A (e.g., and particularly incorporate the base-side (e.g., transmit)wireless power transfer structure 224 shown in FIG. 2A or 2B). Spacesfor two electric vehicles are illustrated in a parking area to be parkedover corresponding base wireless power transfer units 1502 a and 1502 b.In some implementations, a local distribution center 1530 may beconnected to a power backbone 1532 and configured to provide analternating current (AC) or a direct current (DC) supply through a powerlink 1508 to the base wireless power transfer units 1502 a and 1502 b.Each of the base wireless power transfer units 1502 a and 1502 b alsoincludes a base coil 1504 for wirelessly transferring power. The basecoil 1504 may correspond to the coil 226 of the base wireless powertransfer structure 224 as described with reference to FIGS. 2A and 2B.As described above the any of the object detection systems describedabove with respect to FIGS. 1-14 may be integrated into the basewireless power transfer units 1502 a and 1502 b.

The electric vehicle 1512 may include a battery unit 1518, an electricvehicle coil 1516, and an electric vehicle wireless power transfer unit1514. The electric vehicle coil 1516 may correspond to the coil 266 ofthe wireless power transfer structure 260 with reference to FIG. 2B.

The electric vehicle wireless power transfer unit 1514 may receive powerwhen the electric vehicle coil 1516 is located in an electromagneticfield produced by the base coil 1504. The field may correspond to aregion where energy output by the base coil 1504 may be captured by theelectric vehicle coil 1516. For example, the energy output by the basecoil 1504 may be at a level sufficient to charge or power the electricvehicle battery 1518.

In some implementations, the electric vehicle coil 1516 may be alignedwith the base coil 1504 and, therefore, disposed within a near-fieldregion simply by the electric vehicle operator positioning the electricvehicle 1512 such that the electric vehicle coil 1516 is sufficientlyaligned relative to the base coil 1504.

Base wireless charging units (e.g., 1502 a) may be located in a varietyof locations. As non-limiting examples, some suitable locations includea parking area at a home of the electric vehicle 1512 owner, parkingareas reserved for electric vehicle wireless charging modeled afterconventional petroleum-based filling stations, and parking lots at otherlocations such as shopping centers and places of employment.

Charging electric vehicles wirelessly may provide numerous benefits. Forexample, charging may be performed automatically, virtually withoutdriver intervention or manipulation thereby improving convenience to auser. There may also be no exposed electrical contacts and no mechanicalwear out, thereby improving reliability of the wireless charging system1500. Safety may be improved since manipulations with cables andconnectors may not be needed and there may be no cables, plugs, orsockets to be exposed to moisture in an outdoor environment. Inaddition, there may also be no visible or accessible sockets, cables, orplugs, thereby reducing potential vandalism of power charging devices.Further, since the electric vehicle 1512 may be used as distributedstorage devices to stabilize a power grid, a convenient docking-to-gridsolution may help to increase availability of vehicles forvehicle-to-grid (V2G) operation.

FIG. 16 is a schematic view illustrating an example implementation of awireless power transfer system 1600 that may be a portion of thewireless charging system 1500 of FIG. 15. The wireless power transfersystem 1600 of FIG. 16 includes a wireless power transmit circuit 1602.The wireless power transmit circuit includes a transmit resonant circuit1624 including a transmit coil 1608 having an inductance L1. Thetransmit coil 1608 may correspond to the wireless power transfer coil226 with reference to FIG. 2A or 2B.

The wireless power transfer system 1600 further includes a wirelesspower receive circuit 1632. The wireless power receive circuit 1632includes a receive resonant circuit 1642 including a receive coil 1638having an inductance L2. The receive coil 1638 may correspond to thewireless power transfer coil 266 with reference to FIG. 2B.Implementations described herein may use capacitively loaded conductorloops (e.g., multi-turn coils) forming a resonant structure that iscapable of more efficiently coupling energy from a primary structure(transmitter) to a secondary structure (receiver) via a magnetic orelectromagnetic near-field (e.g., where both the transmit resonantcircuit 1624 and the receive resonant circuit 1642 are tunedsubstantially to a common resonant frequency). Using resonant structuresfor coupling energy may be referred to as “magnetically coupledresonance,” “electromagnetically coupled resonance,” and/or “resonantinduction.”

With reference to FIG. 16, a power supply 1622 (e.g., AC or DC) with avoltage Vs supplies power P_(S) to the transmit power conversion circuit1604. In some implementations, the transmit power conversion circuit1604 includes circuitry such as an AC-to-DC converter configured toconvert power from standard mains AC to DC power at a suitable DCvoltage level, and a DC-to-AC frequency converter configured to convertDC power to power at an operating frequency f_(wpt) suitable forwireless high power transfer (e.g., at 85 kHz). In some implementations,the transmit power conversion circuit 1604 includes an inverter amongstother circuitry for regulating the current Is drawn from the powersupply 1622, the current I₁ in the transmit resonant circuit 1642, thevoltage V₁ across the transmit resonant circuit 1624, and the power P₁supplied to the transmit resonant circuit 1624. The transmit resonantcircuit 1624 includes a tuning capacitor 1606 with capacitance C₁ inseries with the transmit coil 1608 to compensate for the reactance ofthe transmit coil 1608 substantially at the operating frequency f_(wpt).When excited by a current I₁, the transmit coil 1608 generates amagnetic field at the operating frequency f_(wpt).

While the transmit resonant circuit 1624 is shown as series-tuned thisshould be considered as just one example implementation. Resonantcircuits 1624 a, 1624 b, and 1624 c provide just a few examples of otherresonant topologies of parallel tuned, parallel series tuned, and thelike. Other implementations may use further reactive components such asfurther capacitors or additional inductors (e.g., for filtering/matchingand other purposes to form other topologies—e.g., LCL-topology, etc.).The transmit coil 1608 receives the power P₁ from the transmit powerconversion circuit 1604 (e.g., AC current is driven into the transmitcoil 1608) and wirelessly transmits power at a level sufficient tocharge or power a load connected to the wireless power receive circuit1632. For example, the level of power provided wirelessly by thetransmit coil 1608 may be on the order of kilowatts (kW) (e.g., anywherefrom 1 kW to 110 kW, although actual levels may be higher or lower).

The receive coil 1638 may be positioned within the near-field of thetransmit coil 1608. In this case, the transmit coil 1608 and the receivecoil 1638 may become wirelessly coupled (e.g., via a magnetic field) toone another such that power may be transferred wirelessly from thetransmit coil 1608 to the receive coil 1638.

The series capacitor 1636 with capacitance C₂ may be provided to form areceive resonant circuit 1642 with the receive coil 1638 having aninductance L₂ that resonates substantially at the operating frequencyf_(wpt). The series-tuned receive resonant circuit 1642 should beconstrued as being exemplary and other resonant topologies may be usedsimilarly as described with the transmit resonant circuit 1624 (e.g.,with respect to resonant circuits 1624 a, 1624 b, and 1624 c to form anycombination of parallel or series topology).

FIG. 16 also indicates magnetic field coupling between transmit coil1608 and receive coil 1638 with a coupling coefficient k(x,y,z) that isa function of the vertical separation (z) and horizontal displacement(x,y). If sufficiently coupled to the transmit coil 1608, the receiveresonant circuit 1642 receives and provides the power P₂ to a receivepower conversion circuit 1634 of the wireless power receive circuit1632. An electrical current in the receive resonant circuit 1642 isrepresented by I₂ and a voltage across the receive resonant circuit 1642is represented by V₂.

The receive power conversion circuit 1634 may include, among otherthings, an AC-to-DC converter configured to convert power at anoperating frequency back to DC power at a voltage V_(L) of the load 1640that may represent a battery unit. In some implementations, the receivepower conversion circuit 1634 includes a rectifier amongst othercircuitry (e.g., DC-DC converter) for regulating the current I₂ in thereceive coil 1638), the voltage V₂ across the receive resonant circuit1642, the current I_(L) and the power P_(L) supplied to the load 1640.

The transmit power conversion circuit 1604 and/or the transmit resonantcircuit 1624 may further include other matching and/or filter circuitryconfigured to match impedances between the transmit resonant circuit1624 and the transmit power conversion circuit 1604 and provide otherfilter functions (e.g., harmonics filtering). Otheramplifiers/oscillators and other circuitry for driving the transmit coil1608 with the appropriate current/voltage is also contemplated.Likewise, the receive power conversion circuit 1634 and/or the receiveresonant circuit 1642 may also include similar matching and/or filtercircuitry.

While the term “coil” is used above, the transmit coil 1608 and thereceive coil 1638 may also correspond to other forms of power transferelements. As such, the term “coil” is intended to refer to a componentthat may wirelessly output or receive energy for coupling to another“coil.” However, in particular exemplary implementations the transmitcoil 1608 and the receive coil 1638 are configured as coils of wire(e.g., Litz wire) and be configured as “conductor loops”, and morespecifically, “multi-turn conductor loops.”

While an example has provided for the implementations described hereinto be part of an electric vehicle charging system, other thesetechniques/implementations described with reference to FIGS. 1-14 may beused for other applications.

The various operations of methods described above may be performed byany suitable means capable of performing the corresponding functions.The means may include various hardware and/or software component(s)and/or module(s), including, but not limited to a circuit, anapplication-specific integrated circuit (ASIC), or processor. Generally,where there are operations illustrated in figures, those operations mayhave corresponding counterpart means-plus-function components withsimilar numbering.

As used herein, the term “determining” encompasses a wide variety ofactions. For example, “determining” may include calculating, computing,processing, deriving, investigating, looking up (e.g., looking up in atable, a database, or another data structure), ascertaining, and thelike. Also, “determining” may include receiving (e.g., receivinginformation), accessing (e.g., accessing data in a memory), and thelike. Also, “determining” may include resolving, selecting, choosing,establishing, and the like.

As used herein, a phrase referring to “at least one of” a list of itemsrefers to any combination of those items, including single members. Asan example, “at least one of: a, b, or c” is intended to cover: a, b, c,a-b, a-c, b-c, and a-b-c, as well as any combination with multiples ofthe same element (e.g., a-a, a-a-a, a-a-b, a-a-c, a-b-b, a-c-c, b-b,b-b-b, b-b-c, c-c, and c-c-c or any other ordering of a, b, and c).

The various illustrative logical blocks, modules and circuits describedin connection with the present disclosure may be implemented orperformed with a general-purpose processor, a digital signal processor(DSP), an ASIC, a field programmable gate array (FPGA) or otherprogrammable logic device (PLD), discrete gate or transistor logic,discrete hardware components, or any combination thereof designed toperform the functions described herein. A processor may be amicroprocessor, but in the alternative, the processor may be anycommercially available processor, controller, microcontroller, or statemachine. A processor may also be implemented as a combination ofcomputing devices, e.g., a combination of a DSP and a microprocessor, aplurality of microprocessors, one or more microprocessors in conjunctionwith a DSP core, or any other such configuration.

The methods disclosed herein comprise one or more steps or actions forachieving the described method. The method steps and/or actions may beinterchanged with one another without departing from the scope of theclaims. In other words, unless a specific order of steps or actions isspecified, the order and/or use of specific steps and/or actions may bemodified without departing from the scope of the claims.

The functions described may be implemented in hardware, software,firmware, or any combination thereof. If implemented in hardware, anexample hardware configuration may comprise a processing system in awireless node. The processing system may be implemented with a busarchitecture. The bus may include any number of interconnecting busesand bridges depending on the specific application of the processingsystem and the overall design constraints. The bus may link togethervarious circuits including a processor, machine-readable media, and abus interface. The bus interface may be used to connect a networkadapter, among other things, to the processing system via the bus. Thebus may also link various other circuits such as timing sources,peripherals, voltage regulators, power management circuits, and thelike.

It is to be understood that the claims are not limited to the preciseconfiguration and components illustrated above. Various modifications,changes and variations may be made in the arrangement, operation anddetails of the methods and apparatus described above without departingfrom the scope of the claims.

What is claimed is:
 1. An apparatus for detecting objects, the apparatuscomprising: a plurality of sense circuits, each of the plurality ofsense circuits comprising: a primary sense coil having a first terminaland a second terminal; a secondary sense coil having a first terminaland a second terminal; and a capacitor having a first terminal and asecond terminal, the first terminal of the capacitor is electricallyconnected to the second terminals of each of the primary sense coil andthe secondary sense coil; a driver circuit electrically connected to thefirst terminal of the primary sense coil of each of the plurality ofsense circuits; and a measurement circuit electrically connected to thefirst terminal of the secondary sense coil of each of the plurality ofsense circuits.
 2. The apparatus of claim 1, further comprising adetection circuit electrically connected to the measurement circuit andconfigured to determine whether an object is proximate the primary sensecoil or the secondary sense coil of at least one of the plurality ofsense circuits based on an output of the measurement circuit.
 3. Theapparatus of claim 2, wherein the detection circuit is configured todetermine whether the object is proximate the primary sense coil or thesecondary sense coil based on determining a magnitude of a change inmutual impedance between the primary sense coil and the secondary sensecoil based on the output of the measurement circuit.
 4. The apparatus ofclaim 1, wherein the driver circuit comprises a current source circuitcomprising: an amplifier circuit having an amplifier output; and amultiplexer electrically connected between the amplifier output and theplurality of sense circuits and configured to selectively connect eachof the plurality of sense circuits to the amplifier output.
 5. Theapparatus of claim 4, wherein the measurement circuit is configured tomeasure a voltage at the first terminal of the secondary sense coil. 6.The apparatus of claim 1, wherein the driver circuit is configured tooperate as a current source and is configured to maintain an outputwherein changes in electrical impedance of each of the plurality ofsense circuits or changes in voltage at the first terminal of thesecondary sense coil have a substantially negligible impact on anelectrical current signal provided by the driver circuit.
 7. Theapparatus of claim 1, wherein each of the primary sense coil and thesecondary sense coil of the plurality of sense circuits is positioned tobe distributed over a predetermined area at least partially defined by awireless power transmit coil configured to inductively transfer power.8. The apparatus of claim 7, wherein a frequency of a magnetic fieldgenerated by the wireless power transmit coil is different than anoperating frequency of a sense signal output by the driver circuit. 9.The apparatus of claim 1, wherein the capacitor is configured tocompensate mutual reactance between the primary sense coil and thesecondary sense coil substantially at an operating frequency of a sensesignal output by the driver circuit.
 10. The apparatus of claim 9,wherein an overall mutual impedance of the primary sense coil and thesecondary sense coil including the capacitor is primarily resistive. 11.The apparatus of claim 1, wherein the primary sense coil is positionedadjacent the secondary sense coil or the primary sense coil is at leastone of partially or wholly overlapping with or interleaved with thesecondary sense coil.
 12. The apparatus of claim 1, wherein the primarysense coil and the secondary sense coil form a bifilar windingstructure.
 13. The apparatus of claim 1, wherein the primary sense coilis magnetically tightly coupled with the secondary sense coil.
 14. Theapparatus of claim 1, further comprising a detection circuit configuredto determine a magnitude of a change in mutual impedance between theprimary sense coil and the secondary sense coil based on an output ofthe measurement circuit.
 15. The apparatus of claim 1, wherein themeasurement circuit comprises an output multiplexer configured toselectively connect each of the plurality of sense circuits to ameasurement amplifier circuit.
 16. The apparatus of claim 1, whereineach of the plurality of sense circuits further comprises a shuntinductor electrically coupled to the first terminal of the secondarysense coil.
 17. The apparatus of claim 16, wherein the capacitor and theshunt inductor are configured as a high pass filter configured toattenuate influence of a wireless power transfer field.
 18. Theapparatus of claim 1, wherein the driver circuit comprises a pluralityof resistors, each of the plurality of resistors respectively connectedto each of the plurality of sense circuits.
 19. The apparatus of claim1, wherein the capacitor is shared between the plurality of sensecircuits, wherein each of the plurality of sense circuits furthercomprises: a first switch electrically connected between the secondterminal of the primary sense coil and ground; and a second switchelectrically connected between the second terminal of the secondarysense coil and ground.
 20. The apparatus of claim 1, wherein themeasurement circuit comprises an amplifier and a feedback resistor, theamplifier having a first input selectively electrically connected eachof the plurality of sense circuits, the feedback resistor electricallyconnected between the first input and an output of the amplifier.
 21. Amethod for measuring changes in electrical characteristics for detectingobjects, the method comprising: applying, from a driver circuit, aninput signal at an operating frequency to a first terminal of a primarysense coil, a second terminal of the primary sense coil electricallyconnected to a first terminal of a capacitor; measuring, at ameasurement circuit, an electrical characteristic at an outputelectrically connected to a first terminal of a secondary sense coil, asecond terminal of the secondary sense coil electrically connected tothe first terminal of the capacitor; and detecting whether an object isproximate to the primary sense coil or the secondary sense coil based onthe electrical characteristic.
 22. The method of claim 21, whereindetecting whether the object is proximate comprises determining amagnitude of a change in mutual impedance between the primary sense coiland the secondary sense coil based on the electrical characteristic. 23.The method of claim 21, wherein the primary sense coil and secondarysense coil form a portion of a sense circuit of a plurality of sensecircuits and wherein the method further comprises sequentially measuringelectrical characteristics of each of the plurality of sense circuits.24. The method of claim 21, wherein the capacitor is configured tocompensate mutual reactance between the primary sense coil and thesecondary sense coil substantially at an operating frequency of theinput signal.
 25. An apparatus for detecting objects, the apparatuscomprising: a plurality of sense circuits, each of the plurality ofsense circuits comprising: a primary sense coil; a secondary sense coil;means for compensating mutual reactance between the primary sense coiland the secondary sense coil substantially at an operating frequency ofa sense signal; means for applying an input signal to each of theplurality of sense circuits; and means for measuring an output signal atan output of the secondary sense coil.
 26. The apparatus of claim 25,further comprising means for detecting whether an object is proximate tothe primary or the secondary sense coil based on the output signal. 27.The apparatus of claim 26, wherein the means for detecting furthercomprises means for determining a magnitude of a change in mutualimpedance between the primary sense coil and the secondary sense coilbased on the output signal.
 28. An apparatus for detecting objects, theapparatus comprising: a plurality of sense circuits, each of theplurality of sense circuits comprising: a primary sense coil; asecondary sense coil; and a capacitor configured to compensate mutualreactance between the primary sense coil and the secondary sense coilsubstantially at an operating frequency of a sense signal; a drivercircuit electrically connected to the primary sense coil of each of theplurality of sense circuits; a measurement circuit electricallyconnected to the secondary sense coil of each of the plurality of sensecircuits; and a detection circuit electrically connected to themeasurement circuit and configured to determine whether an object ispresent based on determining a magnitude of a change in mutual impedancebetween the primary sense coil and the secondary sense coil based on anoutput of the measurement circuit.
 29. The apparatus of claim 28,wherein each of the primary sense coil and the secondary sense coil ofthe plurality of sense circuits is positioned to be distributed over apredetermined area at least partially defined by a wireless powertransmit coil configured to inductively transfer power.
 30. Theapparatus of claim 28, wherein each of the plurality of sense circuitsfurther comprises a shunt inductor electrically coupled to the secondarysense coil.